U.S. patent number 4,282,452 [Application Number 06/020,866] was granted by the patent office on 1981-08-04 for method and circuit arrangement for energizing ultrasonic transducers which are utilized in impulse echo technology.
This patent grant is currently assigned to Siemens Aktiengesellschaft. Invention is credited to Dieter Hassler, Robert Schwarz.
United States Patent |
4,282,452 |
Hassler , et al. |
August 4, 1981 |
Method and circuit arrangement for energizing ultrasonic
transducers which are utilized in impulse echo technology
Abstract
In an exemplary embodiment particularly applicable to the
examination of the human body, for the purpose of energizing with
the aim of transmission of an ultrasonic pulse, a high frequency
pulse of specific duration and amplitude is in each instance
supplied to the ultrasonic transducer. In spite of a substantially
reduced maximum energizing voltage, optimum conditions are to be
created with regard to the intensity of pulses to be radiated with
a simultaneous short pulse duration, so that preferably also
switches which are restricted in their maximum switching voltage
can be utilized for controlling the ultrasonic transducers. This
becomes possible by virtue of the fact that each ultrasonic
transducer is subjected to a specifiable number of periods of a
sinusoidal or sine-like oscillation with a frequency which
energizes the ultrasonic transducer at its useful resonance
frequency. Subsequently thereto, a stop signal, preferably an
additional number of periods of sinusoidal oscillation of smaller
amplitude and with a phase jump of 180.degree. is supplied to the
ultrasonic transducer. After-oscillations of the transducer are
thus extinguished with certainty.
Inventors: |
Hassler; Dieter (Uttenreuth,
DE), Schwarz; Robert (Hessdorf, DE) |
Assignee: |
Siemens Aktiengesellschaft
(Berlin & Munich, DE)
|
Family
ID: |
6035760 |
Appl.
No.: |
06/020,866 |
Filed: |
March 15, 1979 |
Foreign Application Priority Data
|
|
|
|
|
Mar 30, 1978 [DE] |
|
|
2813729 |
|
Current U.S.
Class: |
310/317; 310/326;
367/137 |
Current CPC
Class: |
G01S
15/101 (20130101) |
Current International
Class: |
G01S
15/00 (20060101); G01S 15/10 (20060101); H01L
041/08 () |
Field of
Search: |
;310/316,317,326,327
;73/596-600,606,609-611,629,632 ;128/660,663 ;318/316 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Budd; Mark O.
Attorney, Agent or Firm: Hill, Van Santen, Steadman, Chiara
& Simpson
Claims
We claim as our invention:
1. A circuit arrangement for carrying out the method
comprising:
generating an electric high frequency pulse in the form of a
driving periodic oscillation of predetermined amplitude and of a
driving frequency corresponding to a useful resonance frequency of
an ultrasonic transducer, and having a first phase, generating a
stop signal whose frequency is the same as said driving frequency
but having a second phase offset by substantially 180.degree. in
relation to the first phase of said driving periodic oscillation,
applying to said ultrasonic transducer a predetermined number of
periods of said driving periodic oscillation (7) of said
predetermined amplitude and of said driving frequency and of said
first phase, and then applying to said ultrasonic transducer said
stop signal (9) having the same frequency as said driving
frequency, having said second phase offset by substantially
180.degree. in relation to said first phase, and having a
substantially smaller amplitude than said predetermined amplitude
of said periodic oscillation (7),
said circuit arrangement comprising
an ultrasonic transducer having the useful resonance frequency,
oscillator means for generating a driving periodic oscillation of
said predetermined amplitude and of said driving frequency
corresponding to said useful resonance frequency and of said first
phase, and for thereafter generating said stop signal of the same
frequency as said driving frequency but having a second phase
offset by substantially 180.degree. relative to said first phase,
and
control means controlling said oscillator means to supply to said
ultrasonic transducer first said predetermined number of periods of
said driving periodic oscillation of said predetermined amplitude,
and then said stop signal, such that said stop signal as applied to
said ultrasonic transducer has a substantially smaller amplitude
than said predetermined amplitude of said driving periodic
oscillation,
said oscillator means comprising first means (5) for supplying a
first sinusoidal oscillation (7) as said driving periodic
oscillation, and second means (6, 8) for supplying a second
sinusoidal oscillation (9) as said stop signal with said second
sinusoidal signal having about one-half the amplitude and a
180.degree. phase shift in comparison to said first sinusoidal
oscillation, and said control means comprising switching means (10,
11) between said first and second means and said ultrasonic
transducer for first supplying said predetermined number of periods
of said first sinusoidal oscillation (7) to said ultrasonic
transducer and then supplying to said ultrasonic transducer a
predetermined number of periods of said second sinusoidal
oscillation (9), said second means comprising an output (6)
supplying a sinusoidal oscillation corresponding to said first
sinusoidal oscillation, and phase inversion and amplitude
attenuation means (8) connected with said output (6) and with said
switching means (10, 11) for supplying said second sinusoidal
oscillation (9) to said switching means.
2. A circuit arrangement according to claim 1, with said oscillator
means comprising an oscillator (4) with a quartz
frequency-determining element having a resonant frequency
corresponding to said useful resonance frequency of said ultrasonic
transducer.
3. A circuit arrangement according to claim 1, with said oscillator
means comprising an electrical capacitance and inductance
frequency-determining circuit tuned to said useful resonance
frequency of said ultrasonic transducer.
4. A circuit arrangement according to claim 1, with said oscillator
means comprising a Wien-bridge-oscillator tuned to said useful
resonance frequency of said ultrasonic transducer.
5. A circuit arrangement according to claim 1, with said oscillator
means comprising an RC-phase shifter oscillator tuned to said
useful resonance frequency of said ultrasonic transducer.
6. A circuit arrangement according to claim 1, with said oscillator
means comprising an oscillator with an all-pass network of the
second order tuned to said useful resonance frequency of said
ultrasonic transducer.
7. A circuit arrangement according to claim 1, with said oscillator
means being controlled by said control means to supply to said
ultrasonic transducer about two oscillations of substantially
sinusoidal waveform (7) as said periodic oscillation and for
supplying about one oscillation of substantially sinusoidal
oscillation (9) as said stop signal.
8. A circuit arrangement for carrying out the method
comprising:
generating an electric high frequency pulse in the form of a
driving periodic oscillation of predetermined amplitude and of a
driving frequency corresponding to a useful resonance frequency of
an ultrasonic transducer, and having a first phase, generating a
stop signal whose frequency is the same as said driving frequency
but having a second phase offset by substantially 180.degree. in
relation to the first phase of said driving periodic oscillation,
applying to said ultrasonic transducer a predetermined number of
periods of said driving periodic oscillation (7) of said
predetermined amplitude and of said driving frequency and of said
first phase, and then applying to said ultrasonic transducer said
stop signal (9) having the same frequency as said driving
frequency, having said second phase offset by substantially
180.degree. in relation to said first phase, and having a
substantially smaller amplitude than said predetermined amplitude
of said periodic oscillation (7),
said circuit arrangement comprising
an ultrasonic transducer having the useful resonance frequency,
oscillator means for generating a driving periodic osciallation of
said predetermined amplitude and of said driving frequency
corresponding to said useful resonance frequency and of said first
phase, and for thereafter generating said stop signal of the same
frequency as said driving frequency but having a second phase
offset by substantially 180.degree. relative to said first phase,
and
control means controlling said osicllator means to supply to said
ultrasonic transducer first said predetermined number of periods of
said driving periodic oscillation of said predetermind amplitude,
and then said stop signal, such that said stop signal as applied to
said ultrasonic transducer has a substantially smaller amplitude
than said predetermind amplitude of said driving periodic
oscillation, said oscillator means comprising first and second
triggerable oscillators (19, 20), and said control means (22) being
responsive to supply of a predetermined number of oscillations (7)
from the first oscillator (19) to trigger said second oscillator
(20) to supply to said ultrasonic transducer a predetermined number
of periods of a periodic oscillation as said stop signal,
an adding element (21) having respective inputs connected to said
first and second triggerable oscillators (19, 20) and having an
output connected to said ultrasonic transducer.
9. A circuit arrangement for carrying out the method
comprising:
generating an electric high frequency pulse in the form of a
driving periodic oscillation of predetermined amplitude and of a
driving frequency corresponding to a useful resonance frequency of
an ultrasonic transducer, and having a first phase, generating a
stop signal whose frequency is the same as said driving frequency
but having a second phase offset by substantially 180.degree. in
relation to the first phase of said driving periodic oscillation,
applying to said ultrasonic transducer a predetermined number of
periods of said driving periodic oscillation (7) of said
predetermined amplitude and of said driving frequency and of said
first phase, and then applying to said ultrasonic transducer said
stop signal (9) having the same frequency as said driving
frequency, having said second phase offset by substantially
180.degree. in relation to said first phase, and having a
substantially smaller amplitude than said predetermined amplitude
of said periodic oscillation (7),
said circuit arrangement comprising
an ultrasonic transducer having the useful resonance frequency,
oscillator means for generating a driving periodic oscillation of
said predetermined amplitude and of said driving frequency
corresponding to said useful resonance frequency and of said first
phase, and for thereafter generating said stop signal of the same
frequency as said driving frequency but having a second phase
offset by substantially 180.degree. relative to said first phase,
and
control means controlling said oscillator means to supply to said
ultrasonic transducer first said predetermined number of periods of
said driving periodic oscillation of said predetermined amplitude,
and then said stop signal, such that said stop signal as applied to
said ultrasonic transducer has a substantially smaller amplitude
than said predetermind amplitude of said driving periodic
oscillation,
said oscillator means comprising a single oscillator (23)
responsive to a start pulse to produce a number of oscillations
corresponding to the total number of oscillations of said driving
periodic oscillation and said stop signal to be supplied to said
ultrasonic transducer, and said control means comprising a
switching device (24) having a first input channel connected
directly with said oscillator (23), having a second input channel,
and having an output connected with said ultrasonic transducer,
said oscillator means further comprising amplitude attenuation and
phase inversion means (8) between said oscillator (23) and said
second input channel to supply said stop signal to said ultrasonic
under the control of said switching device.
10. A circuit arrangement for carrying out the method
comprising:
generating an electric high frequency pulse in the form of a
driving periodic oscillation of predetermined amplitude and of a
driving frequency corresponding to a useful resonance frequency of
an ultrasonic transducer, and having a first phase, generating a
stop signal whose frequency is the same as said driving frequency
but having a second phase offset by substantially 180.degree. in
relation to the first phase of said driving periodic oscillation,
applying to said ultrasonic transducer a predetermined number of
periods of said driving periodic oscillation (7) of said
predetermined amplitude and of said driving frequency and of said
first phase, and then applying to said ultrasonic transducer said
stop signal (9) having the same frequency as said driving
frequency, having said second phase offset by substantially
180.degree. in relation to said first phase, and having a
substantially smaller amplitude than said predetermined amplitude
of said periodic oscillation (7),
said circuit arrangement comprising
an ultrasonic transducer having the useful resonance frequency,
oscillator means for generating a driving periodic oscillation of
said predetermined amplitude and of said driving frequency
corresponding to said useful resonance frequency and of said first
phase, and for thereafter generating said stop signal of the same
frequency as said driving frequency but having a second phase
offset by substantially 180.degree. relative to said first phase,
and
control means controlling said oscillator means to supply to said
ultrasonic transducer first said predetermined number of periods of
said driving periodic oscillation of said predetermined amplitude,
and then said stop signal, such that said stop signal as applied to
said ultrasonic transducer has a substantially smaller amplitude
than said predetermined amplitude of said driving periodic
oscillation,
said oscillator means comprising an oscillator (26) with two
control inputs (16, 27) and an output, said oscillator supplying at
its output in response to a start pulse at one control input (16)
said driving periodic oscillation, and said control means being
responsive to the driving periodic oscillation to actuate the other
control input of said oscillator (26) to supply said stop signal at
its output.
11. A circuit arrangement according to claim 10, with said
oscillator means comprising astable oscillator means for supplying
said periodic oscillation (7) and said stop signal (9').
12. A circuit arrangement according to claim 10, with said
oscillator means comprising astable generator means having a
generator output for supplying a rectangular waveform periodic
oscillation, and an integrator connected with the output of said
generator means for supplying a waveform which is the time integral
of said rectangular waveform periodic oscillation to provide said
driving periodic oscillation of said driving frequency for said
ultrasonic transducer.
13. A circuit arrangement according to claim 12, with said
oscillator means further comprising non-linear means connected with
said integrator for conversion of an output triangular waveform
periodic oscillation into a generally sinusoidal oscillation for
supplying said driving periodic oscillation and said stop
signal.
14. A circuit arrangement according to claim 10, with said control
means actuating the other control input of said oscillator (26) to
produce only about one oscillation of said second phase at the
output of said oscillator (26).
15. A circuit arrangement according to claim 10, with said control
means comprising amplitude attentuation means (29) connected with
the output of said oscillator (26) for transmitting the driving
periodic oscillation to said ultrasonic transducer with a first
amplitude and for introducing substantial attenuation in the
transmission of the stop signal to said ultrasonic transducer to
that the stop signal is supplied with a second amplitude
substantially reduced in comparison to said first amplitude.
Description
BACKGROUND OF THE INVENTION
The invention relates to a method and an apparatus for energizing
ultrasonic transducers which are used in impulse echo technology,
in particular, in the examination of the human body, whereby, in
order to energize for the purpose of transmitting an ultrasonic
pulse, there is fed to the ultrasonic transducer, respectively, an
electric high frequency pulse of fixed duration and amplitude.
Ultrasonic transducers of the cited type are utilized particularly
in the so-called B-scan technique, wherein an examination subject,
for example, the human body, is scanned in a linear fashion with
ultrasonic pulses and wherein the echo signals received from the
subject are correspondingly recorded in a linear fashion into an
areal echo-visual image on a recording apparatus (e.g. a cathode
ray oscilloscope). However, the transducers can likewise be those
of the A-scan technique or another scan technique. In the case of
the B-scan technique, the transducer can be a so-called rotational
transducer with a paraboloid reflector. It can likewise also be a
transducer capable of linear displacement, or a pivotal transducer
for e.g. sector-scan. In this category, finally, are also included
transducers of a compound-scan system, and, in particular, also
so-called ultrasonic arrays, wherein a plurality of adjacently
arranged ultrasonic transducers can be energized in chronological
sequence.
In all these applied instances, the electric excitation of the
ultrasonic transducers (in particular, piezo-electric transducers)
proceeds, in the normal instance, by means of a short pulse of very
high amplitude (several 100 V). This type of excitation is the most
effective method with regard to use and electronic outlay as long
as the voltage amplitudes are not subject to any major restriction.
However, such restrictions occur automatically if electronic
switches are to be utilized for the purpose of controlling
ultrasonic transducers, which switches, due to spatial and
functional dimensioning, are restricted in the maximum switching
(or interruption) voltage. A particular technique field is here the
array technology wherein one or more electronic switches must be
allocated to each individual transducer. For reasons of cost and
also for reasons of improved spatial utilization, there is an
interest in the introduction of integrated switches. The maximum
switching (or interruption) voltage of such switches, however, is
generally restricted to approximately 30 to 40 V. The introduction
of switches with such a limited switching (or interruption) voltage
thus leads to a considerable reduction in the voltage amplitude to
be energized; in the case of application (for example, the human
body), this signifies a considerable loss of penetration depth for
the ultrasonic transmission pulses.
SUMMARY OF THE INVENTION
It is the object of the present invention to disclose a means
whereby, in spite of substantially reduced maximum energization
voltage, optimum conditions are created with regard to the
intensity of pulses to be emitted (or radiated), so that preferably
also switches with the above-described properties can be utilized
for the control of ultrasonic transducers.
The object is achieved in accordance with the invention with a
method of the initially cited type in that the ultrasonic
transducer is subjected to a specifiable number of periods of a
sinusoidal or sine-like oscillation; for example, also a triangular
oscillation, with a frequency which energizes the ultrasonic
transducer at its useful (or wanted) resonance frequency, and that,
subsequently thereto, there is supplied to the ultrasonic
transducer a stop signal so that after-oscillations (or vibrations)
of the transducer are extinguished with the occurrence of the stop
signal.
A circuit arrangement for carrying out the method is inventively
characterized by an oscillator for subjecting the ultrasonic
transducer to a specifiable number of periods of a sinusoidal or a
sine-like oscillation with such a frequency which energizes the
ultrasonic transducer at its useful resonance frequency, which
oscillator, subsequent thereto, produces a stop signal, preferably
a further number of periods of sinusoidal or sine-like oscillation
of smaller amplitude and with a phase shift (or jump) of
180.degree..
The construction of conventional ultrasonic transducers
(attenuation member - ultrasonic transducer matching
layer-propagation medium e.g. according to U.S. Pat. No.
3,663,842), which has, in the meantime, become standard, leads to a
relatively narrow-band system measured on the spectrum of pulse
energization. Only a small spectral component of the conventional
wide-band energization pulse is thus converted into the acoustic
useful signal. The greater portion energizes either other vibration
modes or is briefly stored and subsequently fed back again into the
signal source. However, if the ultrasonic transducer is energized
with a signal waveform whose spectrum largely contains only the
desired useful frequency range, then a substantially smaller
energization amplitude leads to the same useful effect. A
narrower-band energization, however, inevitably results in a longer
energization duration. This would lead, without compensating
measures, to a lengthening (or prolonging) of the radiated pressure
pulse to an undesired extent. The present invention, viewed
spectrally, operates with selective energization on the side of the
signal form. By means of subsequent superimposition with the stop
signal, post-oscillations are cancelled. Thus, the abbreviation (or
shortening) of the radiated pressure pulse results. The compensated
sinusoidal energization according to the teaching of the invention
thus combines the useful effect of a narrow-band energization with
the advantage of the brief duration of the pulse energization.
In an advantageous embodiment of the invention, the stop signal
should be at the most precisely as great in amplitude as the actual
energization amplitude of the sinusoidal or sine-like oscillation,.
The stop signal, in addition, should be tuned in its frequency
spectrum in as narrow-band a fashion as possible to the useful
resonance frquency of the oscillator. A voltage pulse of a defined
rise time, decay time, and pulse duration (trapezoidal form) can
serve as the stop signal. In view of a particular narrow-band
property, in a preferred embodiment, however, the stop signal
should be an additional number of periods (preferably one period)
of sinusoidal oscillation of smaller amplitude and with a phase
shift of 180.degree..
Pulse abbreviation through compensation with oppositely proceeding
oscillations (or vibrations) of the ultrasonic transducer is per se
already the subject of the U.S. Pat. No. 2,651,012. From this
publication, a control system for an ultrasonic transducer is prior
knowledge which, in the case of conventional wide-band pulse
energization, after a specifiable time delay, produces a second
oscillation which is intended to be oppositely directed to the
effect of the primary energization pulse. However, it must be noted
that, in the case of a transducer system, the transducer
oscillation exponentially decaying after emission of the primary
energization pulse, is completely undetermined in the number of
oscillations. The number of oscillations is dependent upon a
plurality of parameters, e.g. acoustic characteristic (or surge)
impedance of the sound propagation medium. Related to such a decay
oscillation having an indeterminate number of oscillations and an
indeterminate attenuation characteristic, according to U.S. Pat.
No. 2,651,012, the compensating oscillation which is to be added
with a delay, given an exactly equal attenuation characteristic,
must then also always have the same phase displacement of
180.degree. relative to the decaying oscillations of the primary
energization. However, such conditions can in practice hardly be
achieved or if so, only with a particularly high circuit-technical
outlay. The forming (or shaping) tuning circuits in the two pulse
channels of the circuit arrangement of the U.S. Pat. No. 2,651,012
alone hardly allow a reliable tuning of primary oscillation and
compensating oscillation in the above-desired manner even when
oscillators are introduced having uniformly specified properties. A
change of such an oscillator into one having other properties then,
however, already leads to a complete mistuning of the entire
control system. The change of the oscillator thus requires
re-tuning which, in the case of the described control system of the
aforementioned U.S. patent, must take place on a plurality of
individual tuning members, such as tuning capacitors, tuning
inductances, and tuning resistances. In contrast herewith, in the
present invention, the control of the ultrasonic transducer
proceeds with a fixedly specified number of periods of a sinusoidal
or sine-like oscillation at the useful resonance frequency of the
transducer. However, the duration of the energization of the
transducer at its useful resonance frequency is thus exactly
fixedly specified. At the end of the energization time, the
transducer vibration decays with natural (or self resonant)
frequency without further energization. The stop pulse set
precisely at the end of the energization produces
counter-oscillation of the transducer with, in turn, a fixedly
specified number of periods (preferably one period). This effects a
second decaying operation subsequent to energization by the stop
pulse, again with the resonance frequency of the transducer, but
with opposite phase. Since the transducer in both instances,
specifies the decay properties, no phase displacement (or shift)
can take place between the first decay oscillation, due to primary
excitation, and the second decay oscillation, due to the stop
pulse. The two decay oscillations are independent of the properties
of the transducer, or the stray parameters in the control circuit.
There thus results exact compensation of the decay operations and
hence optimum pulse reduction in the above sense.
In practice, equal echo amplitudes as in the state of the art are
obtained in the manner according to the invention with excitation
(or energization) amplitudes which are already smaller by at least
a factor of five (5) compared with those of the methods of the
state of the art. This already permits the introduction of control
switches with a relatively low switching (or interruption) voltage.
The energizing phase is exactly restricted in duration and
extremely short. There thus results a defined interference-free
echo signal reception. The gain by the factor five results from the
adaptation (or matching) of the excitation waveform to the general
properties (natural or self resonance) specifically of the
ultrasonic transducer. Other major parameters of the transducer
systems utilized today, such as e.g. the acoustic characteristic
(or surge) impedance of the attenuation member and the adaptation
layer, have not yet been taken into consideration. Thus, if these
system parameters are also adapted (or matched) in this manner to
the compensated sinusoidal energization, an additional amplitude
gain by approximately the factor of two (2) results. The
narrow-band compensated sinusoidal energization corresponds more to
the harmonic operation for which the transducer represents a
.lambda./2-line whose terminal impedance is transformed to the
sound emitting acoustic gate (interface). Thus, it is advantageous
if, given a compensated sinusoidal energization, in a further
embodiment of the invention, the acoustic characteristic (or surge)
impedance of an attenuation member, which is disposed on the
surface of the ultrasonic oscillator opposite the radiation
surface, is also matched to the characteristic impedance of the
propagation medium. Through this adaptation (or matching), the
minimum reflection factor in the receiving operation is
simultaneously achieved. Thus, the compensated sinusoidal
energization already brings about a considerable gain in intensity
even in utilization with a conventional system. The adaptation (or
matching) of the conventional system to the compensated sinusoidal
energization brings about an additional gain.
In a further advantageous embodiment of the invention, the
ultrasonic transducer is to be subjected to a total of two periods
of a sinusoidal oscillation which energize the oscillator at its
useful resonance frequency. A third period of sinusoidal
oscillation having a smaller amplitude with a phase shift of
180.degree. is to follow this. The two periods of energization
oscillation and one period of stop oscillation represent an optimum
with regard to the two demands for as great an acoustic
pressure-amplitude as possible, on the one hand, and as small a
pulse duration as possible, on the other hand.
Further advantages and details of the invention shall be apparent
from the following description of exemplary embodiments on the
basis of the accompanying sheets of drawings in conjunction with
the subclaims; and other objects, features and advantages will be
apparent from this detailed disclosure and from the appended
claims.
BRIEF DESCRIPTION OF THE DRAWINGS
FIGS. 1 through 4 show four different embodiments by means of basic
circuit diagrams;
FIG. 5 consisting of FIGS. 5A and 5B shows a detailed
representation of the embodiment according to FIG. 4; and
FIG. 6 shows a diagram of the chronological progressions or
waveforms of the most significant voltages occurring in the circuit
diagram of FIG. 5.
DETAILED DESCRIPTION
In realizing an embodiment of the invention, the ultrasonic
transducer (e.g. piezo-electric transducer) could be directly
included in an oscillator as the frequency-determining element with
the capacitive component of its electric impedance or with its
mechanical oscillation capability. However, it is disadvantageous
that the capacitance of different transducers can vary greatly
given the same resonance frequency. Thus, a tuning of the
oscillator to the respective transducer would be necessary.
More favorable conditions result when a component independent of
the transducer serves as the frequency-determining element.
The basic circuit diagram of FIG. 1 illustrates an ultrasonic
transducer (piezo-electric transducer) 1 which is covered with an
attenuation member on the side facing the radiation surface. The
radiation surface, for the purpose of adaptation to the examination
medium (here a biological tissue), is provided with an
n.multidot..lambda./4-layer 3 (n=1, 3, 5, etc.), which consists of
a material whose characteristic (or surge) impedance corresponds
approximately to the geometric mean of the characteristic impedance
of the examination medium and the attenuation member. Araldite
preferably serves as the material. Multi-stage or constant
adaptation layers, such as are known from transmission theory, can
also be utilized. In order to activate the ultrasonic transducer 1
with sinusoidal oscillations, there is an oscillator 4 which
comprises, as the frequency determining element, e.g. an oscillator
quartz (or piezo-electric crystal) or electric LC-oscillatory (or
resonant) circuits. Oscillator 4 is free-running; i.e. it
continuously produces at its outputs 5 and 6 a sinusoidal voltage 7
with the natural (or self resonant) oscillation frequency of the
ultrasonic transducer 1. The sinusoidal voltage at the output 6 of
oscillator 4 is shifted in phase through 180.degree. by means of a
phase inversion (or reversal)--and attenuation installation 8, and
is simultaneously attenuated to approximately half the amplitude.
(The amplitude attenuation of apparatus 8 is variably adjustable.)
The result is then the continuous sinusoidal oscillation 9 at the
output of the installation 8. Since both sinusoidal oscillations 7
and 9 are continuous, the desired periods must be cut out
(selected) by the corresponding switching times of analog switches
10 and 11, of a switching installation 12. The actuation (or
control) of switches 10 and 11 in the desired manner proceeds by
means of control logic 13 via control lines 14 and 15,
respectively. The control logic 13 is constructed in such a manner
that, subsequent to the start pulse at the start pulse input 16, it
recognizes, via an interrogation (or sample) line 17 from the
oscillator 4, the next-following passage through zero of the
sinusoidal oscillation 7, and closes switch 11 in the illustrated
switching position of switch 10 via switching line 15. After four
additional passages through zero of the sinusoidal oscillation 7
(end of the second period) the changeover switch 10 is then
operated into the lower switching position by the logic 13 via the
switching line 14. Thus, a switchover operation is carried out from
sinusoidal oscillation 7 to the sinusoidal oscillation 9. Following
termination of a full period of this sinusoidal voltage 9 (third
passage through zero, or first repetition of the passage through
zero in the direction e.g. of negative amplitude) switch 11 is
opened again by logic 13 via the control line 15. There results, as
the product of this control mechanism, the sinusoidal output
oscillation 18 which is composed of a total of two periods of the
sinusoidal oscillation 7 and a period of the sinusoidal oscillation
9 following with a 180.degree. phase shift. In the exemplary
embodiment of FIG. 1, start pulses at the input 16 for the logic 13
are generally the clock pulses of that particular clock pulse
generator (not illustrated) which are emitted in the transmit pulse
of the ultrasonic transmission.
If the oscillator 4 is only to be briefly switched to oscillation
operation during the transmission phases (abandonment of continuous
operation), the frequency-determining elements of the oscillator 4
must manifest short buildup- and decay-times. However, the buildup
and decay is characterized by the transitional behavior of the
frequency-determining element. It can therefore be advantageous, in
such an applied instance, to introduce, as the
frequency-determining element, a relatively wide-band and therefore
rapidly building-up frequency element. Oscillators which meet these
conditions are e.g. the Wien bridge oscillator, the
RC-phase-shifter oscillator, and the oscillator with an all-pass
network of the second order. A further possibility is that, with
the switching on and off, a Dirac-like pulse is switched (or
offered) to the frequency-determining element. Such a pulse
accelerates the buildup and brakes the decay. Basically, it is also
possible to operate (or control) all the described oscillator types
in such a manner that they are switched to transmission in a
preparatory manner a specified period prior to the emission of the
transmit pulse, respectively. The full transmission amplitude is
then already available at oscillator 4 for the respective following
transmission time. Subsequent to release of a transmit signal, the
oscillator can then again be switched off.
By way of contrast, an oscillator which is relatively simple and
problem-free in construction is obtained using a square wave
generator as the basis. Square wave generators can be readily
conceived such that they start to oscillate virtually without delay
(the initial value conditions are relatively simple to adhere to).
By means of integrators, the square wave can be converted into a
triangle. From the triangle it is possible to generate a sine
signal without all too great an outlay, for example, through
introduction of non-linear elements and low-passes, or the like.
This sine signal is hardly distorted (can be practically free of
distortion). The demands regarding the deviation from the pure sine
form as well as the demands for frequency stability, side-band
noise, etc., then correspond in advance to the conventional demands
in pulse-echo technology.
However, integrators which are realizable in practice are never
free of zero point drift. Thus, for this reason, the square wave
signal of the square wave generator should not be directly
integrated. On the contrary, a triangle (or delta) controlled
square wave generator should be introduced whose control delta
voltage is conveyed out of the square wave generator and is
transformed (or converted) to the sine in the described manner.
Accordingly, a frequency-determining element is thus selected whose
transitional behavior approaches as closely as possible the desired
sine progression in the switch-on moment. The integrator is such an
element. Since, however, a sine oscillator with only one integrator
cannot alone be brought to oscillate, but a square wave generator
can be brought to oscillate, and the square wave, in turn, well
corresponds to the step function of the switching-on and -off, in a
preferred embodiment, an astable oscillator is advisable as the
basic element. An astable oscillator as the triggerable function
generator (it supplies the triangle- or delta- and square
wave-signal) can in response to a logic signal, be started and/or
stopped in the passage through zero.
The sample embodiments of FIGS. 2 through 4 operate with astable
oscillators as function generators.
FIG. 2 again illustrates the ultrasonic transducer 1 with
attenuation member 2 and adaptation (or matching) layer 3. The
circuit arrangement for operating the transducer 1 now comprises a
total of two triggerable oscillators 19, 20, and adding element 21,
and a logic control circuit 22. In response to a start pulse at
start input 16, the oscillator 19 emits a double period of the
sinusoidal oscillation 7 which directly reaches the transducer 1
via adding element 21. The end of this double oscillation 7 is
recognized by the logic 22 (either from the passages through zero
or from oscillator-internal signals). The logic 22 subsequently
starts the second triggerable oscillator 20 which supplies a period
of sinusoidal oscillation 9 of opposite phase position and
approximately half the amplitude to the transducer 1 via the adding
element 21. Thus, there again results, relative to transducer 1,
the oscillation combination 18 as it is already illustrated in the
embodiment of FIG. 1.
The exemplary embodiment of FIG. 3 illustrates a modification of
such a type that only a single triggerable oscillator 23 is
introduced which, in response to a start pulse at the input 16,
releases a total of three periods of the sinusoidal oscillation 7.
The first two periods of the sinusoidal oscillation travel directly
in the direction of ultrasonic transducer 1 via a switch 24 in the
illustrated upper switching position. The end of the second period
is, in turn, recognized by a logic circuit 25 which subsequently
switches switch 24 into the lower switching position. In this lower
switching position, the oscillator signal 9, inverted and halved in
amplitude at installation 8, now reaches transducer 1 with the
third and last period. Thus, there again results the desired
composite signal 18. Subsequent to connection of this third
component, at the latest after a new start signal at the input 16
of oscillator 23, switch 24 must again be returned to the
illustrated initial upper switching position.
FIG. 4 illustrates by means of a basic circuit diagram an
embodiment comprising an oscillator 26 which manifests two trigger
inputs 16 and 27. In the case of a start pulse at input 16,
oscillator 26 initially produces at its output two periods of the
sinusoidal voltage 7. The end of the second period is recognized by
the logic circuit 28 and is responded to with a second start pulse
for the input 27 of oscillator 26. Oscillator 26 then produces a
third period oppositely disposed in phase. As illustrated in FIG.
4, the third period, as sinusoidal voltage 9', can manifest the
same amplitude as sinusoidal voltage 7. In order to arrive from
such an oscillation to half the amplitude, the period 9' must then
be adjusted to half the amplitude by means of control amplifier 29.
The control proceeds by means of the logic circuit 28 via the
control input 30 of the control amplifier 29 for the purpose of
amplification factor control. However, oscillator 26 can likewise
also be so designed that, in the case of a start pulse at the start
input 27, it produces a third period of sinusoidal oscillation of
half the amplitude as well as of opposite phase. This oscillation
then would correspond as to amplitude as well as to phase to the
oscillation 9 of the embodiments of FIGS. 1 through 3. In such an
instance, the control amplifier 29 retains its normal
amplification.
A more detailed implementaion according to the teachings of FIG. 4
is illustrated in the circuit diagram of FIG. 5. The mode of
operation of the circuit arrangement of FIG. 5 is shown by the
voltage characteristic waveforms of FIG. 6 and the embodiment of
FIG. 5 is further explained as follows:
In FIG. 5, 26 (FIG. 5A, the upper circuit module) again designates
the oscillator (triggerable delta or triangular waveform
generator); 29 (FIG. 5B) designates the control amplifier (and
output power amplifier); and 28 (FIG. 5A, the lower circuit module)
designates the logic control. In the idle state, transistor T1 in
the oscillator 26 is conductive, so that the input B of the
comparator IC1 (with differential amplifier DV and logic elements
L1 and L2) is held at e.g. an input voltage level of one-half the
supply voltage U.sub.01 or 2.5 volts (U.sub.01 /2=2.5 V). The
integrator, consisting of T2, T3, C2-C7, and R6-R15 is thereby
connected with its output C likewise to e.g. 2.5 V. With the
falling slope of the start signal P (FIG. 6) at the input (FIG. 5A)
of the sequence control logic 28, an approximately 50 ns-long pulse
at P3 is produced which sets to zero the counter Z (with flip-flop
FF1, FF2 and logic elements L4, L5) and the memory SP (with logic
element L3 and inverter I1), so that circuit point E (at the output
of logic element L4, FIG. 5A) goes to logical one (e.g. U.sub.01
=+5 V) and blocks T1. Simultaneously the output B of the comparator
IC1 is forced to logical one. Since the integrator (output C)
cannot follow rapid voltage changes, (the output B of logic element
L2 (FIG. 5A) is, however, coupled back to input B of the
comparator), the output A of the comparator remains in the initial
logical zero condition (0 V); the output B of the comparator IC1
remains at logical one level even after the pulse at P3 has
disappeared. The logic zero at the output A of the comparator IC1
blocks T2 and shifts T3 into the conductive state. T3 now operates
as a constant current source controlling current flow to C3 (and
C4, in case switch S1 for lower frequencies is closed). Since
C5>>C3, C4, the current is similarly transmitted to C2 and
likewise recharges this capacitor. The voltage at point C now
increases in a time-linear fashion until it reaches the voltage
value connected to input B of differential amplifier DV (for
example, +5 V). A slight exceeding of this value is sufficient in
order to cause the comparator IC1 to flip over (trigger); the
voltage at point B at the output of logic element L2 then drops to
enlarge the differential voltage between input A of differential
amplifier DV and input B thereto in the sense of a positive (or
regenerative) feedback until the stable condition output A at
logical one level, output B at logical zero (A=1, B=0) has been
obtained. The integrator operates in an opposite direction (T2
conductive, T3 blocked) until reaching the lower transition (or
switchover) point (C=0 V) etc. The following falling slopes of the
output A (see waveform A of FIG. 6) are jointly counted in counter
Z. At 1/4th period prior to the end of the second full oscillation,
the counter is at the position FF1 set, FF2 reset (1, 0). With the
zero crossing of the a.c. component of waveform C, FIG. 6, at the
end of two complete oscillations, point F goes to logical one, so
that D drops to logical zero. The comparator input B is thereby
brought from logical one to logical zero, and the oscillation is
thereby thrown into the opposite phase position. F thus again drops
and again releases the omparator input B with circuit point D at a
logic one level (D=1). The memory SP is simultaneously set via T6
and I4, and the latter memory ensures that the amplification of the
terminal amplifier drops to e.g. half. Again 1/4 period prior to
the end of the third full oscillation, with the rise of A, E goes
to logical zero and clamps the potential of the comparator input B
at e.g. 2.5 V. The integrator output C can now continue to drop
only to this voltage and remains there until the next start pulse.
In the long pulse pauses the capacitor C5 has the opportunity of
regenerating its charge.
At potentiometer R6 the frequency can be adjusted within narrow
boundaries (i.e. over a narrow range). With potentiometer R11,
small asymmetries of the triangular (or delta) waveform (FIG. 6 at
C) as compared with the zero line can be corrected. With
potentiometer R13, the commencement of the third period can be
varied within narrow boundaries. At terminals P6, P7, identical
a.c. current signals are available with different d.c. voltages, so
that the coupling of a complementary input of the output power
amplifier 29 is unproblematical. The collector resistance of this
stage consists, during the first two periods, of R33 in cooperation
with R32, Z2, Z3. The resistance R33 alone would result in a
voltage amplitude of more than e.g. .+-.20 V. Through the
series-connection of R32, Z2, Z3, the delta (or triangular) peak is
strongly flattened, so that the sine-form is well approximated. The
capacitances of the circuit (particularly of Z2, Z3, FIG. 5B)
perform an extra function for the purpose of rounding the edges by
means of low-pass action (or effect). For the third period, T16,
T17 (FIG. 5B), are rendered conductive, so that R34, R35 and R36
connect (or hook up) and approximately halve (or cut in half) the
collector resistance, as a consequence of which the amplification,
or the output amplitude, respectively, is halved (adjustment
through R34). It is now possible to dispense with a "rounding",
because no blocking voltage problem is present. The transistors T10
through T15 form the power stage. Their transverse current is
automatically adjusted in terms of d.c. current through negative
feedback by means of resistances R30, R31, R41, R43, such that
transfer distortions (in the range of small voltages) remain
sufficiently small. The desired combination oscillation 18
according to FIG. 6 is then available at the output (taken from the
common circuit point between R48 and R49) of the output power
amplifier 29, FIG. 5B.
It will be apparent that many modifications and variations may be
effected without departing from the scope of the novel concepts and
teachings of the present invention.
* * * * *