U.S. patent number 3,835,421 [Application Number 05/401,554] was granted by the patent office on 1974-09-10 for microwave transmission line and devices using multiple coplanar conductors.
This patent grant is currently assigned to RCA Corporation. Invention is credited to Robert Eugene De Brecht, Louis Sebastian Napoli.
United States Patent |
3,835,421 |
De Brecht , et al. |
September 10, 1974 |
MICROWAVE TRANSMISSION LINE AND DEVICES USING MULTIPLE COPLANAR
CONDUCTORS
Abstract
Three coplanar conductive surfaces on the top surface of a
dielectric substrate form a microwave transmission line having
first and second transmission modes used in the construction and
operation of various microwave devices, such as amplifiers
unbalanced-to balanced transmission line transformers and
directional couplers.
Inventors: |
De Brecht; Robert Eugene
(Cranbury, NJ), Napoli; Louis Sebastian (Hamilton Square,
NJ) |
Assignee: |
RCA Corporation (New York,
NY)
|
Family
ID: |
26979708 |
Appl.
No.: |
05/401,554 |
Filed: |
September 27, 1973 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
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315087 |
Dec 14, 1972 |
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Current U.S.
Class: |
333/26;
333/246 |
Current CPC
Class: |
H01P
5/18 (20130101) |
Current International
Class: |
H01P
5/18 (20060101); H01P 5/16 (20060101); H01p
005/10 (); H03h 007/42 () |
Field of
Search: |
;333/26,84M |
Other References
nishide et al., Balance-to-Unbalance Transformers, Monogr. Res.
Inst. Appl. Elec. (Japan), No. 18, (1970), pp. 69-73 relied
on..
|
Primary Examiner: Gensler; Paul L.
Attorney, Agent or Firm: Norton; Edward J. Lazar; Joseph D.
Mahoney; Donald E.
Parent Case Text
This is a division of application Ser. No. 315,087, filed Dec. 14,
1972, now U.S. Pat. No. 3,798,575.
Claims
What is claimed is:
1. An unbalanced-to-balanced transmission line transformer
operative over a desired band of frequencies for transforming a
first impedance to a second impedance comprising:
a dielectric substrate having a predetermined dielectric
constant;
a transmission line for electromagnetic energy, said transmission
line having first, second and third coplanar strip-like conductors
each having input and output ends and predetermined widths and
lengths adjacent to one surface of said dielectric substrate, said
first conductor having one longitudinal edge substantially parallel
to and separated from an adjacent longitudinal edge of said second
conductor by a first predetermined gap, said first conductor having
a second edge opposite and substantially parallel to said one edge,
said second edge being substantially parallel to and separated from
an adjacent longitudinal edge of said third conductor by a second
predetermined gap, said first conductor input end being adjacent to
said second and third conductor input ends and said first conductor
output end being adjacent to said second and third conductor output
ends, said second conductor being arranged to be at a first R.F.
potential relative to said first conductor and said third conductor
being arranged to be at a second different R.F. potential relative
to said first conductor in the presence of said electromagnetic
energy intermediate said ends, said first, second and third
conductor widths, said dielectric substrate, and said first and
second predetermined gaps being arranged to form said transmission
line,
means for establishing a predetermined D.C. potential at both said
second conductor input end and said third conductor input end,
whereby said first, second and third conductor input ends, said
dielectric constant and said first and second predetermined gaps
form at said first, second and third conductor input ends an
unbalanced transmission line input terminal section having said
first impedance; and
means for connecting said second conductor output end to said first
conductor output end, whereby said first and second conductor
output connected ends, said third conductor output end, said
dielectric constant and said first and second predetermined gaps
form at said first, second and third conductor output ends a
balanced transmission line input terminal section having said
second impedance.
2. An unbalanced-to-balanced transmission line transformer
according to claim 1, wherein each of said first, second and third
conductor predetermined lengths from said input to output ends is
substantially .lambda./4, where .lambda. is the wavelength at the
center frequency of said desired frequency band.
Description
DESCRIPTION OF THE PRIOR ART
Existing microwave transmission lines suitable for microwave
integrated circuits employ a strip-like conductor on the top
surface of a dielectric substrate and a ground planar conductor on
the bottom surface of the dielectric substrate. Microwave energy is
confined substantially within the dielectric substrate and is
transmitted from an input port to an output port in the TEM
(transverse electromagnetic) mode. Some microwave integrated
circuits employ two adjacent and coplanar strip-like conductors on
the top surface of the dielectric substrate. The microwave
transmission characteristics of these circuits are dependent on how
the electric fields of the microwave energy are distributed between
conductive surfaces on both sides of the substrate.
For certain devices, it is inconvenient and impractical to use a
transmission line having a ground planar conductor on the bottom
surface of a dielectric substrate and one or two coplanar
strip-like conductors on the top surface of the dielectric
substrate. A prior art transmission line described in U.S. Pat. No.
3,560,893 issued to Cheng Paul Wen on Feb. 2, 1971, describes the
use of three coplanar and parallel strip-like conductors on the top
surface of a dielectric substrate. Microwave energy is transmitted
along the three conductor transmission lines in a first
transmission mode that confines the electric field of the applied
microwave energy between the center conductor and the two outer
ground potential conductors. Certain microwave devices require not
only the first transmission mode but a new second transmission mode
that confines the field between the two outer conductors for
efficient operation.
SUMMARY OF THE INVENTION
According to the present invention a transmission line for
electromagnetic energy comprising first, second and third coplanar
strip-like conductors having predetermined widths adjacent to one
surface of a dielectric substrate confine the electric fields of
the electromagnetic energy substantially within the dielectric
substrate in first and second transmission modes. The first
conductor has a first edge separated from an adjacent edge of the
second conductor at a first relative electric potential by a first
predetermined gap. The first conductor also has a second edge,
opposite the first edge, that is separated from an adjacent edge of
the third conductor at a second relative electric potential by a
second predetermined gap. The first, second and third conductor
widths, the dielectric constant of the dielectric substrate, and
the first and second predetermined gaps are arranged to confine the
electric fields of the electromagnetic energy substantially within
the dielectric substrate between the first and second conductors
and between the first and third conductors in a first transmission
mode and between the second and third conductors in a second
transmission mode.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a perspective view of a prior art electromagnetic energy
transmission line.
FIG. 2 is a perspective view of a transmission line in accordance
with one embodiment of the present invention.
FIG. 3 is a plot of the even mode impedance of the transmission
line shown in FIG. 2 as a function of the ratio of the transmission
line dimensions, a.sub.1 /b.sub.1, the dielectric constant of the
dielectric substrate and the ratio of the transmission line
dimensions c.sub.1 -b.sub.1 /2a.sub.1.
FIG. 4 is a plot of the ratio of odd mode impedance to even mode
impedance as a function of the ratio of the transmission line
dimensions a.sub.1 /b.sub.1, the dielectric constant of the
dielectric substrate and the ratio of the transmission line
dimensions c.sub.1 -b.sub.1 /2a.sub.1.
FIG. 5 is a perspective view of a coplanar conductor directional
coupler in accordance with another embodiment of the present
invention.
FIG. 6 is a perspective view of an unbalanced-to-balanced
transmission line transformer in accordance with another embodiment
of the present invention.
FIG. 7 is a schematic representation of the unbalanced-to-balanced
transmission line transformer illustrated in FIG. 6.
FIG. 8 is a top view of a microwave transistor push-pull amplifier
in accordance with a still further embodiment of the present
invention.
DESCRIPTION OF PREFERRED EMBODIMENTS OF THE INVENTION
Referring to FIG. 1, there is shown a perspective view of a prior
art electromagnetic energy transmission line. The transmission line
comprises three coplanar strip-like conductors 11, 15 and 17 on the
top surface 22 of a dielectric substrate 13. The prior art
transmission line, illustrated in FIG. 1, is described in U.S. Pat.
No. 3,560,893 issued to Cheng Paul Wen on Feb. 2, 1971. A single,
thin relatively narrow strip-like conductor 11 is separated by
predetermined gaps from two relatively wide strip-like conductors
15 and 17 both at the same R.F. and D.C. ground potential. The
minimum width of the first relatively wide strip-like ground
conductor 15 is more than twice as wide as narrow strip-like
conductor 11 and is spaced near to and parallel with coplanar
narrow strip-like conductor 11. The minimum width of the second
relatively wide strip-like ground conductor 17 is likewise more
than twice as wide as narrow strip-like conductor 11 and is spaced
near to and parallel with coplanar narrow strip-like conductor 11
but on the opposite side of narrow strip-like conductor 11 relative
to ground conductor 15. The top surface 22 of dielectric substrate
13 having the three coplanar strip-like conductors 11, 15 and 17
thereon is open to free space. The bottom surface 16 of dielectric
substrate 13 is likewise open to free space.
The distribution of the electric field of electromagnetic energy
coupled to the transmission line from a source, not shown, is
represented by dashed electric field lines 19. Electric field lines
19 are distributed only between narrow strip-like conductor 11 and
wider strip-like conductors 15 and 17 along their entire lengths.
Electric field lines 19 are contained mainly within the dielectric
substrate 13 between narrow strip-like conductor 11 and ground
conductors 15 and 17. However, some electric field lines 19, not
shown, are distributed between conductor 11 and conductors 15 and
17 in the free space region. The intensity of the electric field
within dielectric substrate 13 is dependent on the magnitude of the
dielectric constant of dielectric substrate 13. A discontinuity in
displacement current density at the interface between dielectric
substrate 13 and free space is produced by that portion of electric
field 19 tangential to the air-dielectric boundary on dielectric
surface 22. The discontinuity in displacement current on dielectric
surface 22 produces an axial component of magnetic field,
represented by dashed lines 20, associated with electric field 19.
A portion of the axial component of the magnetic field 20 at the
interface between dielectric substrate 13 and free space on surface
22 is in the direction of propagation. The magnetic field 20
extends along both sides of narrow conductor 11 and passes under
narrow conductor 11. The distance, d, between conductors 15 and 17
is preferably less than one-half wavelength at the operating
frequency in order to prevent the transmission of electromagnetic
energy in undesired modes.
In order to distribute the electric field between narrow strip-like
conductor 11 and wider strip-like conductors 15 and 17, a
difference in potential must exist between narrow strip-like
conductor 11 and wider strip-like conductors 15 and 17. Since both
wider strip-like conductors 15 and 17 are at the same R.F. and D.C.
ground potential, the boundary conditions for establishing an
electric field between conductors 15 and 17 does not exist. The
characteristic impedance of the prior art three coplanar strip-like
conductor transmission line is dependent on the establishment of an
electric field between only narrow strip-like conductor 11 and
wider strip-like conductors 15 and 17.
Referring to FIG. 2, there is shown a perspective view of a
transmission line 40 comprising three coplanar strip-like
conductors 21, 25 and 27 on the top surface 35 of a dielectric
substrate 23 in accordance with one embodiment of the present
invention. Center strip-like conductor 21 is separated by
predetermined gaps from first and second outer strip-like
conductors, 25 and 27 having predetermined widths. First outer
strip-like conductor 25 is spaced near to and parallel with
coplanar center strip-like conductor 21. Second outer strip-like
conductor 27 is spaced near to and parallel with coplanar center
strip-like conductor 21 but on the opposite side of center
strip-like conductor 21 relative to first outer conductor 25. The
top surface 35 of dielectric substrate 23 having the three coplanar
strip-like conductors 21, 25 and 27 thereon as illustrated in FIG.
2 is open to free space. Unlike the prior art transmission line
illustrated in FIG. 1, the minimum width of first and second outer
conductors 25 and 27 according to the present invention is not
limited to be at least twice as wide as center conductor 27.
Contrary to the prior art arrangements, according to the present
invention a difference in R.F. potential is provided between outer
conductors 25 and 27 in accordance with several arrangements to be
described.
Electromagnetic energy from a source, not shown, is coupled to
transmission line 40. The distribution of the electric field of the
electromagnetic energy coupled to transmission line 40 is
represented by dashed electric field lines 29. Electromagnetic
energy can be transmitted along transmission line 40 in a first
transmission mode that distributes electric field lines 29 between
center strip-like conductor 21 and outer strip-like conductors 25
and 27 along their entire lengths since a difference in R.F.
potential exists between center strip-like conductor 21 and outer
strip-like conductors 25 and 27. Electromagnetic energy can also be
transmitted along transmission line 40 in a second transmission
mode that distributes electric field lines 29 between only outer
strip-like conductors 25 and 27. Conditions can be established as
will be apparent to those skilled in this art that would allow
simultaneous transmission of electromagnetic energy in both the
first and second transmission modes. It should be understood that
the terms "first" and "second" transmission modes designate for
convenience of description and the appended claims both the
arrangements and the modes of operation of transmission line 40
according to the present invention.
A portion of electric field 29 is tangential to the air-dielectric
boundary on dielectric surface 35 and produces a discontinuity in
displacement current density at the interface between dielectric
substrate 23 and free space. The discontinuity in displacement
current on dielectric surface 35 produces an axial component of
magnetic field, represented by dashed lines 31, associated with
electric field 29. A portion of the axial component of the magnetic
field 31 at the interface between dielectric substrate 23 and free
space on surface 35 is in the direction of propagation when the
distance, d, between outer conductors 25 and 27 is small compared
to the electrical distance of one wavelength at the operating
frequency. Under these conditions, the magnetic field lines 31
extend along both sides of center conductor 21 and eventually pass
under center conductor 21 forming a closed magnetic loop having a
portion in the direction of electromagnetic transmission. The
magnitude of the magnetic field 31 present at the air-dielectric
interface on one side of center conductor 21 is not equal to the
magnitude of the magnetic field 31 on the other side of center
conductor 31. If the magnetic field lines 31 were represented by
magnetic field vectors, the vectors would appear at the
air-dielectric interface on the top surface of dielectric substrate
23 between center conductor 21 and outer conductors 25 and 27. The
vectors would have a magnitude and direction that would define a
condition of circular polarization existing on the top surface of
dielectric substrate 23 between center conductor 21 and outer
conductors 25 and 27. The sense of circular polarization (clockwise
or counterclockwise) would be the same if viewed on opposite sides
of center conductor 21. This polarization condition is significant
in the construction of microwave ferrite devices as described by
Lax and Button in Chapter 12 of "Microwave Ferrites and
Ferrimagnetics," McGraw-Hill publication.
If the distance, d, between outer conductors 25 and 27 is larger
than the electrical distance of one wavelength at the operating
frequency, the axial component of magnetic field lines 31 form a
closed loop around center conductor 21. Under this condition, the
closed loop of magnetic field lines 31 around center conductor 21
is then transverse to the direction of electromagnetic
transmission.
The term even mode impedance, Z.sub.OE, is used to identify the
impedance of transmission line 40 when electromagnetic energy is
transmitted in the first transmission mode, viz., when electric
field lines 29 are distributed between center strip-like conductor
21 and outer strip-like conductors 25 and 27. The impedance
Z.sub.OE , refers to an even mode impedance having a magnitude
dependent on the intensity of the electric field distribution
between center conductor 21 and outer conductor 27. The impedance
Z.sub.OE refers to an even mode impedance having a magnitude
dependent on the intensity of the electric field distribution
between center conductor 21 and outer conductor 25.
Referring to FIG. 3, there is shown a graph of even mode impedance,
Z.sub.OE, in terms of the relative dielectric constant,
.epsilon..sub.r, of dielectric substrate 23 versus the ratio of the
substrate dimensions, a.sub.1 /b.sub.1, shown in FIG. 2. The
dimension a.sub.1 is the distance from the center line of center
conductor 21 to the edge of center conductor 21. The dimension
b.sub.1 is the distance from the center line of center conductor 21
to the nearest edge of outer conductor 25. The dimension c.sub.1 is
the distance from the center line of center conductor 21 to the
furthest edge of outer conductor 25. Assuming that the dimensions
of strip-like conductors 21, 25 and 27 comprising transmission line
40 and the relative dielectric constant .epsilon..sub.r are known,
the graph in FIG. 3 is useful for determining even mode impedance
Z.sub.OE under the condition that transmission line 40 is
symmetrical or Z.sub.OE = Z.sub.OE and the substrate thickness, t,
greater than 4 .times. b.sub.1. In other words, the dimensions
b.sub.1 and c.sub.1 determining the width of outer conductor 25 and
the gap between center conductor 21 and outer conductor 25 in FIG.
2, also correspond to the dimensions of the width of outer
conductor 27 and the gap between center conductor 21 and outer
conductor 27. The magnitude of the even mode impedance is
independent of the thickness, t, of dielectric substrate 23 in FIG.
2 when thickness, t, exceeds 4 .times. b.sub.1.
The term odd mode impedance, Z.sub.OO, is used to identify the
impedance of transmission line 40 when it is transmitting
electromagnetic energy in the second transmission mode, viz., when
electric field lines 29 are distributed between only outer
conductors 25 and 27.
Referring to FIG. 4, there is shown a graph of the ratio of odd
mode impedance to even mode impedance, Z.sub.OO /Z.sub.OE, versus
the ratio of the substrate dimensions, a.sub.1 /b.sub.1, shown in
FIG. 2. The graph in FIG. 4 is useful for determining odd mode
impedance, Z.sub.OO, knowing Z.sub.OE and the dimensions of
strip-like conductors 21, 25 and 27 comprising transmission line 40
provided either transmission line 40 is symmetrical or Z.sub.OE =
Z.sub.OE . As explained above, transmission line 40 is symmetrical
when the dimensions b.sub.1 and c.sub.1 determining the width of
outer conductor 25 and the gap between center conductor 21 and
outer conductor 25 in FIG. 2 also correspond to the dimensions of
the width of outer conductor 27 and the gap between center
conductor 21 and outer conductor 27. The magnitude of the odd mode
impedance is independent of the thickness, t, of the dielectric
substrate 23 in FIG. 2 when thickness, t, exceeds 4 .times.
b.sub.1.
FIGS. 3 and 4 illustrate that the even mode and odd mode impedances
of transmission line 40 in FIG. 2 vary as a function of the a.sub.1
/b.sub.1 ratio when transmission line 40 has a predetermined ratio
of outer conductor width to center conductor width (c.sub.1
-b.sub.1 /2a.sub.1). A transmission line for electromagnetic energy
comprising three coplanar strip-like conductors on the top surface
of a dielectric substrate permits construction of passive devices
requiring a determination of even and odd mode impedances for
improved operation. The disclosed transmission line configuration
permits easy connection of active devices between center conductor
21 and outer conductors 25 and 27 as well as a similar connection
of other passive components.
Referring to FIG. 5, there is shown a perspective view of a
directional coupler, according to this invention, comprising three
coplanar strip-like conductors 51, 55 and 57 on the top surface 65
of a dielectric substrate 53. A directional coupler is a passive
microwave device used for dividing microwave energy coupled to an
input port between two output ports. The propagation of microwave
energy transmitted to each output port is dependent on the desired
coupling coefficient. Part of the energy reflected at the two
output ports is directed to a fourth port usually terminated in an
energy absorbing load. The design of a directional coupler in terms
of even mode, Z'.sub.OE, and odd mode, Z'.sub.OO, impedances is
known being described by Matthaei, Young and Jones in Chapter 13 of
"Microwave Filters Impedance-Matching Networks, and Coupling
Structures." The desired characteristics of a directional coupler
(coupling coefficient, bandwidth, etc.) are directional coupler
design goals described in Chapter 13 of the above cited text and
are used to calculate the magnitude of Z'.sub.OE and Z'.sub.OO. The
magnitude Z.sub.OE used in FIGS. 3 and 4 is equivalent to Z'.sub.OE
/2. The magnitude of Z.sub.OO used in FIGS. 3 and 4 is equivalent
to 2Z'.sub.OO. Thus, FIGS. 3 and 4 can be used to determine the
widths of conductors 51, 55 and 57 and the separation between
center conductor 51 and outer conductors 55 and 57 that would allow
operation of a directional coupler having desired operating
characteristics.
FIG. 5 also illustrates a method of coupling microwave energy to
and from a transmission line comprising three coplanar strip-like
conductors 51, 55 and 57 on the top surface 65 of dielectric
substrate 53. Coaxial outer conductor 58 of coaxial connector 66 is
connected to outer strip-like conductor 57. Coaxial center
conductor 70 of connector 66 is connected to the closest end of
center strip-like conductor 51. Coaxial outer conductor 59 of
coaxial connector 67 is connected to outer strip-like conductor 57.
Coaxial center conductor 71 of connector 67 is connected to the
closest end of center strip-like conductor 51. A length L.sub.1 of
outer strip-like conductor 57 separates coaxial outer conductor 58
of connector 66 from coaxial outer conductor 59 of connector 67.
The length L.sub.1 is equivalent to an electrical length of
substantially .lambda./4, where .lambda. is the wavelength
determined by the equation:
.lambda. = C/f.sqroot..epsilon..sub.r +1/2 (1)
C being the velocity of light in a vacuum, f the mid-band operating
frequency and .epsilon..sub.r the relative dielectric constant of
dielectric substrate 53.
Coaxial outer conductor 61 of coaxial connector 69 is connected to
outer strip-like conductor 55. Coaxial center conductor 73 of
connector 69 is connected to center strip-like conductor 51 at the
same end as coaxial center conductor 70 of coaxial connector 66.
Coaxial outer conductor 60 of connector 68 is connected to outer
strip-like conductor 55. Coaxial center conductor 72 of connector
68 is connected to center strip-like conductor 51 at the same end
as coaxial center conductor 71 of connector 67. A length L.sub.3 of
outer strip-like conductor 55 separates coaxial outer conductor 61
of connector 69 from coaxial outer conductor 60 of connector 68.
The length L.sub.3 is equivalent to an electrical length of
substantially .lambda./4, where .lambda. is the wavelength
determined by equation (1).
Outer strip-like conductor 57 may be at the same D.C. potential as
outer strip-like conductor 55 if either coaxial outer conductors 58
or 59 is at the same D.C. potential as either of coaxial outer
conductors 60 or 61. However, outer strip-like conductors 55 and 57
are not at the same R.F. potential when coaxial outer conductors
58, 59, 60 and 61 are connected to outer strip-like conductors 55
and 57 as illustrated in FIG. 5. Thus, by arranging the connections
as just described, the previously discussed boundary conditions for
exciting the even and odd mode impedances in a transmission line
comprising three coplanar strip-like conductors are preserved.
The length L.sub.2 of center strip-like conductor 51 is equivalent
to an electrical length of .lambda./4, where .lambda. is the
wavelength determined by equation (1). Center strip-like conductor
51 is coextensive and parallel with outer strip-like conductor 55
over length L.sub.3 with outer strip-like conductor 57 over length
L.sub.1.
It is well known that a microwave signal coupled to an input port
connector of a directional coupler may be divided into two output
signals that are coupled from two output port connectors that are
directly opposite the input port connector. For example, if a
microwave signal is coupled to input port connector 66, part of the
microwave signal is transmitted directly to directly opposite
output port connector 67 and part of the microwave signal is
coupled to directly opposite output port connector 69.
Substantially none of the input microwave signal is coupled to
diagonally opposite connector 68.
Referring to FIG. 6, there is shown according to this invention a
perspective view of an unbalanced-to-balanced transmission line
transformer commonly referred to as a balun. The balun provides an
impedance transformation from the impedance magnitude of the signal
source, not shown, coupled to the unbalanced transmission line
input terminal section 74 to the impedance magnitude of a load, not
shown, coupled to the balanced transmission line output terminal
section 78. The balanced transmission line output terminal section
78 consists of two coplanar strip-like conductors 79 and 80 on the
top surface 95 of dielectric substrate 75. The balun is designed to
transmit energy to a load terminating conductors 79 and 80. The
design of balun section 125 determines the characteristic impedance
of balanced transmission line 78 and section 125 also provides a
condition that establishes a phase difference of 180 electrical
degrees between conductors 79 and 80.
Unbalanced transmission line input terminal section 74 consists of
an arrangement of three coplanar and parallel strip-like conductors
81, 85 and 87 in the top surface 95 of dielectric substrate 75 more
fully described in U.S. Pat. No. 3,560,893 issued to C. P. Wen on
Feb. 2, 1971. Relatively narrow strip-like center conductor 81 is
separated by predetermined gaps from two relatively wide strip-like
conductors 85 and 87 both at the same R.F. and D.C. ground
potential. One method of establishing the same R.F. and D.C. ground
potential at conductors 85 and 87 is to connect the outer conductor
of a coaxial connector, not shown, to conductors 85 and 87 and the
center conductor of the connector to conductor 81. The width, W, of
outer conductors 85 and 87 is at least twice as wide as the width
of center conductor 81. A length of 10 mil diameter wire 76 is
connected from outer conductor 85 to outer conductor 87. Wire 76 is
used to maintain the same R.F. and D.C. ground potential between
outer strip-like conductors 85 and 87 at the end of unbalanced
transmission line input terminal section 74.
Balun section 125 consists of three coplanar and parallel
strip-like conductors 81, 88 and 89 on the top surface 95 of
dielectric substrate 75. Center strip-like conductor 81 is
separated by predetermined gaps from outer strip-like conductors 88
and 89. One end of outer strip-like conductor 88 is connected to
outer strip-like conductor 87 near the connection point of wire 76.
The other end of conductor 88 is connected to one end of strip-like
conductor 77. The electrical length of conductor 88 from the
connection point of wire 76 to the connection point of conductor 77
is substantially .lambda./4, where .lambda. is the wavelength
determined by equation (1). The electrical length of conductor 77
is negligible. One end of center strip-like conductor 81 is
connected to the other end of strip-like conductor 77. One end of
outer strip-like conductor 89 is connected to outer strip-like
conductor 85 near the connection point of wire 76.
One end of strip-like conductor 79 of section 78 is connected to
conductor 77 anywhere along the length of conductor 77. One end of
strip-like conductor 80 of section 78 is illustrated in FIG. 6 as
being an extension of outer strip-like conductor 89. Such an
arrangement is exemplary only of other possible arrangements. The
end of strip-like conductor 80 may be connected to conductor 89
anywhere along the end of conductor 89.
Referring to FIG. 7, there is shown a schematic equivalent of
section 125 of the balun illustrated in FIG. 6. The impedance
Z.sub.O is the load impedance terminating the balanced transmission
line output terminal section 78. The schematic representation of
section 125 is useful in explaining the determination of odd mode
impedance, Z.sub.OO, and even mode impedance, Z.sub.OE, necessary
for the design of a balun operative over a broad frequency band.
Section 125 is schematically illustrated as having a first short
circuited transmission line stub section 100, having a
characteristic impedance 2Z.sub.OE, connected in shunt with a
transmission line section 102 having a characteristic impedance
2Z.sub.OE. A second short circuited transmission line stub section
101 having a characteristic impedance Z.sub.OO is also connected in
shunt with transmission line section 102. The electrical length of
transmission line section 102 separating the connection points of
sections 100 and 101 to section 102 is substantially .lambda./4,
where .lambda. is the wavelength defined by equation (1). The
electrical length of sections 100 and 101 from their connection to
section 102 to their short circuited ends is substantially
.lambda./4, where .lambda. is the wavelength defined in equation
(1).
The characteristic impedance, Z.sub.O ', of balanced transmission
line section 78 at mid-band frequency, f.sub.o, is
Z.sub.O ' = (2Z.sub.OE).sup.2 /Z.sub.O (2)
where Z.sub.OE is the even mode impedance of section 102 and
Z.sub.O is the magnitude of the impedance of the signal source, not
shown, coupled to unbalanced transmission line section 74. At
mid-band frequency, f.sub.o, the shunt connected short circuited
stub sections 100 and 101 each appear as an open circuit or very
high impedance connected in shunt with section 102 and thus do not
affect the determination of balanced transmission line
characteristic impedance Z.sub.O '.
At operating frequencies other than the mid-band frequency,
f.sub.o, the characteristic impedance Z.sub.bal of balanced
transmission line section 78 is:
1/Z.sub.O = j(1/2Z.sub.OE +1/Z.sub.OO) cot .theta.+(sec.sup.2
/2Z.sub.OE Z.sub.O - j2cot.theta.)(1/2Z.sub.OE) (3)
where Z.sub.OE is the even mode impedance of section 102, Z.sub.OO
is the odd mode impedance of section 102, .theta. is the electrical
length of stub sections 100 and 101 at the operating frequency and
Z.sub.O is the magnitude of the signal source, not shown, coupled
to unbalanced transmission line section 74. Thus, the desired
characteristic impedance Z.sub.O ' of balanced output terminal
section 78 and its variations over a desired frequency band can be
determined from equations (2) and (3). The even mode impedance,
Z.sub.OE, and the odd mode impedance, Z.sub.OO, used in equations
(2) and (3) together with the graphs of FIGS. 3 and 4 may be used
to determine the width of strip-like conductors 81, 88 and 89 and
the spacing between center conductor 81 and outer conductors 88 and
89 of the balun illustrated in FIG. 6.
Referring to FIG. 8, there is shown a top view of a microwave
transistor push-pull amplifier according to the invention, having
all conductive surfaces and transistors on the top surface 95 of a
dielectric substrate. The push-pull amplifier uses the balun
illustrated in FIG. 6 as push-pull amplifier input transformer 103
and push-pull amplifier output transformer 104. For convenience,
the numbers identifying the conductive surfaces of the balun
illustrated in FIG. 6 are used to identify the conductive surfaces
of input and output push-pull amplifier transformers 103 and 104. A
detailed explanation of push-pull transformer-coupled power
amplifiers is disclosed in Section 4.2 of "Electronic Designers'
Handbook" by Landee, Davis and Albrecht.
Gate electrode 105 of transistor T.sub.1 is connected to balanced
transmission line terminal 80 of input balun 103 and gate electrode
106 of transistor T.sub.2 is connected to balanced transmission
line terminal 79 of input balun 103. Source electrode 107 of
transistor T.sub.1 and source electrode 108 of transistor T.sub.2
are connected to strip-like conductors 109 which are at D.C. ground
potential. As previously discussed, in the description of the balun
illustrated in FIG. 6, the widths of strip-like conductors 81, 88
and 89 and the separation between inner conductor 81 and outer
conductors 88 and 89 of input balun 103 are determined from FIGS. 3
and 4 when the magnitudes of even mode impedance, Z.sub.OE, and odd
mode impedance, Z.sub.OO, are known. Equations (2) and (3) are used
to determine the magnitudes of Z.sub.OE and Z.sub.OO necessary for
the proper impedance transformation from the known impedance of the
input signal source, not shown, to the known input impedance
magnitude of transistors T.sub.1 and T.sub.2. As an example, the
impedance of the input signal source is 50 ohms and the magnitude
of the combined input impedance of transistors T.sub.1 and T.sub.2
is substantially 200 ohms. The relative dielectric constant,
.epsilon..sub.r, of the dielectric substrate is 2.2. The width of
center strip-like conductor 81 of balun 103 is 0.020 inches. The
widths of outer strip-like conductors 88 and 89 of balun 103 is
0.020 inches. The separation between center strip-like conductor 81
and outer strip-like conductors 88 and 89 of balun 103 is 0.028
inches.
Drain electrode 110 of transistor T.sub.1 is connected to balanced
transmission line terminal 80 of output balun 104 and drain
electrode 126 of transistor T.sub.2 is connected to balanced
transmission line terminal 79 of output balun 104. The widths of
strip-like conductors 81, 88 and 89 and the separation between
inner conductor 81 and outer conductors 88 and 89 of output balun
104 are determined from FIGS. 3 and 4 when the magnitudes of even
mode impedance, Z.sub.OE, and odd mode impedance, Z.sub.OO, are
known. Equations (2) and (3) are used to determine the magnitudes
of Z.sub.OE and Z.sub.OO necessary for the proper impedance
transformation from the output impedance magnitude of transistors
T.sub.1 and T.sub.2 to the impedance magnitude of the load, not
shown, terminating the output signal port. As an example, the
impedance of the terminating output load is 50 ohms and the
magnitude of the combined output impedance of transistors T.sub.1
and T.sub.2 is substantially 450 ohms. The width of center
strip-like conductor 81 of balun 104 is .016 inches. The widths of
outer strip-like conductors 88 and 89 of balun 104 is .016 inches.
The separation between center strip-like conductor 81 and outer
strip-like conductors 88 and 89 of balun 104 is .035 inches.
A negative D.C. bias voltage of 2 volts is applied to gates 105 and
106 of Gallium Arsenide Schottky-barrier FET (Field Effect
Transistor) transistors T.sub.1 and T.sub.2. A positive D.C. bias
voltage of 5 volts is applied to drains 110 and 126 of transistors
T.sub.1 and T.sub.2. The gain of the push-pull amplifier was 1.5 db
over a 1.0 GHz band of frequencies centered at 5.2 GHz. The
magnitude of the output power was 20 mw and the efficiency of the
amplifier was 13 percent.
* * * * *