Crystal Filters

Abstract

Three or more resonator-forming electrode pairs and a crystal wafer on which they are mounted, form a multiresonator crystal filter with respective inductors connected to two of the pairs. The electrode pairs have masses such as to tune the frequency exhibited by the unconnected resonator to a frequency f.sub.p. The inductors tune the interelectrode capacitances of the connected resonators to the frequency f.sub.p. The masses of electrodes in the connected resonators tune the mechanical resonance of the crystal between the electrodes to the frequency f.sub.p. The electrode spacings in view of their masses are such as to achieve predetermined couplings between resonators.

Description

REFERENCE TO RELATED APPLICATIONS

This application relates to the copending applications Ser. No. 541,549, filed Apr. 11, 1966 and Ser. No. 558,338, filed June 17, 1966, both of W. D. Beaver and R. A. Sykes and assigned to the assignee of the present application. This application also relates to the copending application of R. L. Reynolds and R. A. Sykes Case 1--20, filed on or about Mar. 30, 1968 and also assigned to the assignee of this application. The subject matter of these applications are herewith made a part of this application as if included herein.

BACKGROUND OF THE INVENTION

This invention relates to energy transfer devices using the acoustical resonant properties of crystals, and particularly to electric wave filters using an acoustically resonant crystal body having mounted thereon separate electrode pairs to form respective mutually coupled resonators and wherein electrical energy applied to one of the electrode pairs is coupled through the body and removed from another so that the filter transmits a predetermined passband having predetermined characteristics.

While some kind of energy transmission can be expected whenever electrical energy is applied to a pair of electrodes piezoelectrically coupled to a crystal body and energy removed from another pair of electrodes on the body, it has not always been possible to obtain a controlled preselected characteristic or a controlled smooth passband.

It has been proposed that the passbands be controlled with additional components. Also the before-mentioned copending applications disclose that controlling the masses and spacing of electrode pairs allow the character of the transmission characteristic or passband to be controlled without additional components as long as the passband was limited to a frequency range less than the frequency difference between the so-called resonant and antiresonant frequencies of one electrode pair. It was also discovered that within this frequency range the passband could be controlled by mounting a number of extra electrode pairs on the body between the further-separated input electrode pair and output electrode pair. These intermediate electrode pairs served with the body to form resonators that coupled the input resonator, formed by the input electrodes and the crystal body, to the output resonator, formed by the output electrodes and the crystal body. However, such additional resonators were not suitable outside of that resonant-to-antiresonant frequency range. In structures where an intermediate electrode pair served to couple the input and output pairs, the interelectrode capacitances of each electrode pair imposed restrictions on the passband, which if they could be overcome at all, appeared to require such cumbersome changes as to forbid practical application. Thus such energy translating devices or crystal filters possess practical limitations.

THE INVENTION

According to a feature of the invention three or more resonator-forming electrode means covering respective portions of the crystal body so that one of the electrode means is acoustically coupled through the crystal body to the other two. Furthermore, the other electrode means are given masses and dimensions such that each portion of the body covered by the other two or more electrode means exhibits a predetermined uncoupled mechanical resonant frequency. Also respective inductances tune the capacitances formed by the electrodes of the other electrode means to the predetermined resonant frequency. At the same time the first electrode means are made sufficiently massive to form with the portion of the body which it covers a resonator having a tuning frequency in a given relation with the mechanical resonant frequency.

According to another feature of the invention this given relation is such that the tuning frequency is equal to the predetermined mechanical resonant frequency. According to still another feature of the invention the mechanical resonant frequency of each of the other electrode means is the same.

According to yet another feature of the invention the electrode means which couples the other two electrode means comprises a pair of short-circuited electrodes. According to still another feature of the invention the one electrode means which couples the other two electrode means comprises a pair of open-circuited electrodes. In the case of the short-circuited electrodes the uncoupled mechanical resonant frequency of the body covered by one electrode pair preferably is equal to the predetermined mechanical resonant frequency of the other electrode means.

By virtue of these features the passband limits previously imposed by the interelectrode capacitances are eliminated with only two inductors and without the need of additional reactive components connected to the intermediate electrodes.

These and other features of the invention are pointed out in the claims. Other objects and advantages of the invention will become obvious from the following detailed description when read in light of the accompanying drawing.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a partly schematic, partly plan drawing of a circuit including a filter with a crystal body and embodying features of the invention;

FIG. 2 is a partly schematic, partly sectional drawing of the circuit in FIG. 1;

FIG. 3 is a schematic drawing of an all-electrical equivalent of the circuit in FIGS. 1 and 2;

FIG. 4 is a graph illustrating the transmission characteristic of the filter in FIGS. 1 and 2;

FIG. 5 is a partly schematic, partly sectional drawing of a test arrangement for testing couplings between resonators in the circuit of FIGS. 1 and 2;

FIG. 6 is a partly schematic, partly sectional drawing of another circuit embodying features of the invention; and

FIGS. 7, 8 and 9 are drawings illustrating characteristics useful for manufacturing the filters in the circuits of FIGS. 1, 2 and 6.

DESCRIPTION OF PREFERRED EMBODIMENTS

In FIG. 1 a source S applies a high frequency potential across a tuning inductor L.sub.1 and across the first of eight pairs of electrodes 12, 14; 16, 18; 20, 22; 24, 26; 28, 30; 32, 34; 36, 38; and 40, 42, that are vapor-deposited or plated in alignment along a chosen axis, such as the Z' crystallographic axis on a rectangular AT-cut quartz crystal wafer or body 44. For clarity, the thicknesses of the electrodes and wafer in FIG. 1 are exaggerated. The individual electrodes of each of the pairs oppose each other across the wafer. The source S by applying the high frequency potential across the input electrodes 12 and 14 piezoelectrically generates thickness shear vibrations in the crystal wafer 44. The vibrations excite vibrations of equal frequency in the crystal wafer portions between successive pairs of electrodes 12 to 42 and generate electrical energy in the electrodes 40 and 42 across which a tuning inductor L.sub.2 appears. Each electrode pair with the wafer forms a resonator coupled to the adjacent resonators. A load resistor R.sub.o receives electrical energy appearing across the output electrodes 40 and 42 over a predetermined bandwidth B.sub.w. The intermediate pairs of electrodes 16 through 38 are all short circuited to each other and grounded.

The masses of the electrodes 12 through 42 affect the resonant frequency of each of the resonators when considered alone by lowering their respective frequencies from the fundamental thickness shear mode frequency of the crystal wafer 44. Each of the masses is such as to tune the individual resonator in the absence of the others to the same frequency f.sub.O when measured in the short circuit condition. The inductor L.sub.i tunes the total effective electrical interelectrode capacitance C.sub.01 of the first electrode pair 12 and 14, including the stray capacitances, to the resonant frequency f.sub.O so that f.sub.0 equals 1/(2.pi. L.sub.i C.sub.01). The inductor L.sub.o tunes the interelectrode capacitance of the last pair of electrodes 40, 42 to the same frequency f.sub.0 so that f.sub.0 equals 1/(2.pi. L.sub.o C.sub.02). In FIG. 1 the electrodes 12 through 42 are substantially equal so that C.sub.01 = C.sub.08 = C.sub.0, and L.sub.i = L.sub.o.

The operation of the filter in FIG. 1 may best be understood by considering it in connection with the equivalent circuit shown in FIG. 3 wherein the portions representing the structure composed of the crystal body 44 and the electrodes 12 through 42 are designated F. Here, the source S applies high frequency potentials across a capacitor C.sub.01 representing the electrical interelectrode capacitance of the electrodes 12 and 14. The source S has an internal resistance R.sub.s. A capacitance transformer composed of a capacitance Tee having two capacitor C.sub.1 in the series arm and a capacitor -C.sub.1 in the shunt leg represent the piezoelectric coupling between the electrodes 12 and 14 and the wafer 44 and serve to apply the input energy from the source S to a shunt resonant circuit composed of an inductor L.sub.1 and capacitor C.sub.1 representing the resonant structure of the wafer 40 between the electrodes 12 and 14. The resonant circuit composed of inductor L.sub.1 and C.sub.1 is tuned to the frequency f.sub.0 on the basis of the thickness of wafer 44, the dimensions and masses of electrodes 12 and 14. Thus, 1/(2.pi. C.sub.1 L.sub.1)= f.sub.0. The inductor L.sub.i connected across the capacitor C.sub.01 and the electrodes 12 and 14 tunes the capacitor C.sub.01 to the frequency f.sub.0 so that the latter equals 1/(2.pi. L.sub.i C.sub.01 ). The energy appearing in the resonator L.sub.1 C.sub.1 excites a plurality of resonators L.sub.2, C.sub.2 ; L.sub.3, C.sub.3 ; L.sub.4, C.sub.4 ; L.sub.5, C.sub.5 ; L.sub.6, C.sub.6 ; L.sub.7, C.sub.7 ; and L.sub.8, C.sub.8. Here C.sub.1 = C.sub.2 = C.sub.3 = C.sub.4 = C.sub.5, = C.sub.6 = C.sub.7 = C.sub.8, and L.sub.1 = L.sub.2 = L.sub.3 = L.sub.4 = L.sub.5 = L.sub.6 = L.sub.7 = L.sub.8. This is done by means of inductive .pi. coupling sections S.sub.12, S.sub.23, S.sub.34, S.sub.45, S.sub.56, S.sub.67 and S.sub.78, each of which is composed of a series inductor L.sub.n(n+1) and two shunt arms -L.sub.n(n+1), where n = 1, 2...7. Thus the section S.sub.34 is composed of a series inductor L.sub.34 and two shunt inductors -L.sub.34. The excitation of the other resonators corresponds to the effect of the vibration of wafer 44 between electrodes 12 and 14 exciting through the intermediate material similar vibrations of the same frequency in the wafer portions between the respective pairs of electrodes 16 to 42. The respective resonators L.sub.1, C.sub.1, etc., each represent the vibration frequency of the individual resonating portion of the wafer when the other resonating portions are detuned from the passband region. The inductive sections S.sub.12, S.sub.23, etc., represent the coupling between successive resonators and result in frequency shifts of the vibrating portions between the wafers.

The energy appearing across the resonator L.sub.8, C.sub.8, representing the energy between the electrodes 40 and 42, is piezoelectrically coupled to the capacitance C.sub.08 representing the total interelectrode capacitance across the electrodes 40 and 42, by means of an output capacitance transformer corresponding to the piezoelectric coupling between the electrodes and wafer 44. A capacitance Tee circuit formed of two series capacitor arms C.sub.1 and C.sub.1 and a shunt leg -C.sub.1, similar to the input capacitance transformer, forms the output capacitance transformer. The electrical energy appearing there passes to a load resistor R.sub.o. The inductor L.sub.o tunes the capacitance C.sub.08 to the frequency f.sub.0.

In the equivalent circuit of FIG. 3, the portion W.sub.e represents the equivalent circuit of the wafer material. The portion T.sub.e represents the piezoelectric coupling between the wafer material and the electrodes and the portion E.sub.e represents the total electrical capacitances of the electrodes.

Corresponding to the conditions of FIG. 1, the electrodes 16 through 38 are short circuited. Therefore, the capacitances C.sub.02 to C.sub.07 are also short circuited. The effect of these short circuits is actually to place an infinite impedance or open circuit across each one of the resonators C.sub.2, L.sub.2 to C.sub.2, L.sub.7. This can be seen from computing the values of the total capacitances C.sub.t across one of the capacitances such as C.sub.2. Here the capacitor C.sub.1 in the leg of the capacitance transformer circuit to C.sub.02 is shunted across the capacitor -C.sub.1. The capacitor -C.sub.1 on the other hand is then in series with the capacitor C.sub.1 in the other arm of the capacitance Tee. The total value of capacitance C.sub.t across the resonator L.sub.2, C.sub.2 equals C.sub.1 (C.sub.1 - C.sub.1)/(C.sub.1 + C.sub.1 - C.sub.1)= 0. The capacitance thus equals zero and the corresponding reactance X.sub.CT = 1/2.pi.fC.sub.t is infinite. Therefore the effect of the short circuits across each of resonators L.sub.21, C.sub.02 to L.sub.7, C.sub.07 is effectively to place infinite reactances across the particular resonators and have substantially no effect upon their tuning.

At the same time the inductors L.sub.i and L.sub.o and the impedances R.sub.s and R.sub.o which form shunt resonant circuits with the capacitors C.sub.01 and C.sub.08 represent respective impedances composed of infinite reactances and respective resistances R.sub.s and R.sub.o at the frequency f.sub.0. Thus at frequency f.sub.o the capacitance Tees reflect these low resistances and parallel-resonant circuits L.sub.i, C.sub.01 and L.sub.o, C.sub.08 across resonators C.sub.1, L.sub.1 and C.sub.8, L.sub.8 as high resistances and zero reactances if the values of R.sub.s and R.sub.o are low. This is so because the reflected impedance such as Z.sub.01 across the resonator L.sub.1, C.sub.1 at f.sub.o equals [(R.sub.s + X.sub.C1)(- X.sub.C1)/(R.sub.s + X.sub.C1 - X.sub.C1)] -X.sub.C1 = X.sub.C1 .sup.2 /R.sub.s, where X.sub.C1 = 1/j2.pi.f.sub.o C.sub.1 is the reactance of C.sub.1 at the frequency f.sub.o. Thus at frequency f.sub.o the reflected impedance Z.sub.01 = 1/4.pi..sup.2 f.sub.o .sup.2 C.sub.1 .sup.2 R.sub.s, a real value. If R.sub.s is small compared to the reactance of C.sub.1 Z.sub.01 is a large resistance. Similarly if R.sub.o is small compared to the reactance of C.sub.1 Z.sub.08 is large.

As a result, each of the resonators C.sub.1, L.sub.1 to C.sub.8, L.sub.8, representing the resonators formed by the electrodes 12 to 42 and the wafer 44 is tuned to substantially the same frequency f.sub.o.

The passband formed by such tuning depends upon the coupling between each successive pair of resonators. FIG. 4 illustrates a passband available from a circuit such as shown in FIGS. 1 and 2. The desired degree of coupling necessary to produce particular passbands is available from ordinary circuit theory. The actual coupling between adjacent resonators can be measured by determining the frequency shifts imparted by one of a pair of resonators upon the other.

The circuit of FIG. 5 illustrates the method for determining the coupling between successive pairs of electrodes. Here, a variable frequency source 60 applies a high frequency signal across one of the two pairs of electrodes between which the coupling is to be measured. A meter 62 measures the input voltage. The resonator to which the coupling is to be measured such as that formed by the electrodes 24 to 26 is short circuited. The remaining electrodes are maintained open circuited to detune the resonators formed by them. The applied frequency from the source 60 is noted at the two lowest voltages measured by the meter 62 as the frequency output of the source 60 varies. These two noted frequencies f.sub.A and f.sub.B are formed by the effect of inductive section S.sub.34 upon the resonators L.sub.3, C.sub.3 and L.sub.4, C.sub.4 which represent the resonators formed by the electrodes 20, 22 and 24, 26. The coupling k is equal to (f.sub.b - f.sub.A)/ f.sub.B f.sub.A. If open circuiting the electrodes, such as 12 to 18 and 28 to 42 in FIG. 5, whose couplings are not being considered in any measurement does not adequately detune them so that they are outside of the measurement scope, additional inductance or capacitance may be added to detune them further.

Suitable dimensions for the structure of FIGS. 1 and 2 follow. These dimensions are only examples and should not be taken as limiting. According to this example the crystal body is composed of an AT-cut quartz crystal 1 inch long, 0.400 inches wide and approximately 0.0061 inches thick. The dimensions of the electrode pairs 12 through 42 are 0.0734 inches along the long direction of the crystal body, that is along the Z' axis, by 0.0916 inches along the X-axis. The electrode separations d.sub.1 through d.sub.7 between the edges having the long dimensions are:

d.sub.1 = 0.01171 inches

d.sub.2 = 0.01478 inches

d.sub.3 = 0.01531 inches

d.sub.4 = 0.01542 inches

d.sub.5 = 0.01531 inches

d.sub.6 = 0.01478 inches

d.sub.7 = 0.01171 inches

These spacing dimensions have tolerances of .+-.0.0001 inches, respectively. The masses of the electrodes are such as to achieve respective platebacks of 2 percent. The term "plateback" is defined in the before-mentioned copending applications and represents a measure of the masses or the effects of the masses of the electrodes. Specifically, plateback constitutes the fractional drop (f- f.sub.0)/f in the resonant frequency f.sub.0 of a crystal body electroded with a single pair of electrodes, from the fundamental thickness shear frequency f of the unelectroded crystal body due to increasing masses of the electrodes. This takes into account the fact that as the masses of the electrodes are increased, the resonant frequency of the individual resonator as measured with other resonators detuned is lowered.

The resulting coupling coefficients k between successive pairs from left to right in FIGS. 1 and 2 are 1.54.times. 10.sup..sup.-3, 1.245.times. 10.sup..sup.-3, 1.200.times. 10.sup..sup.-3, 1.192.times. 10.sup..sup.-3, 1.200.times. 10.sup..sup.-3, 1.245.times. 10.sup..sup.-3 and 1.54.times. 10.sup..sup.-3. The structure of FIGS. 1 and 2 passes the midband frequency of 10.7 megahertz. The width of the passband is 25 kilohertz. The resonator inductance L.sub.1 through L.sub.8 is 34 millihenries. The filter is intended to operate between a source having an impedance of 3000 ohms and a load of 3000 ohms. The inductors L.sub.i and L.sub.o are 0.154 millihenries, respectively. R.sub.s = 3000 ohms.

The frequency bandwidth over which a filter such as shown in FIGS. 1 and 2 operates to furnish a continuous passband as shown in FIG. 4 is wider than hitherto available with such crystal structures. In the past it had been assumed that such smooth bandwidths, rather than being limited by other considerations such as the mechanical coupling between resonators, were limited by the effect on the passband of the piezoelectric coupling of the body to the capacitance of each of the electrodes 12 to 42. However, the equivalent circuit of FIG. 3, reveals that the piezoelectric coupling operates to impose the interelectrode capacitance differently upon the intermediate resonators than on the input and output resonators. Specifically when the source and load are applied across the input and output resonators the capacitors C.sub.1 in the arms of the capacitor Tee lie in the main energy path. This has a different effect from the capacitor Tee on the intermediate resonators C.sub.2, L.sub.2 to C.sub.7, L.sub.7. There the capacitor Tee representing the piezoelectric coupling is substantially shunted across the energy path. This effect is true even when the electrodes 16 to 38 are open circuited.

While short circuiting of intermediate electrodes 16 through 38 is often desirable for the purpose of limiting the effects upon the capacitance C.sub.o, that is any of the capacitances C.sub.01 to C.sub.08 of leads and reflected capacitances, the open-circuited condition is often desirable. Under these circumstances, the equivalent circuit of FIG. 3 changes only by having the short circuits of C.sub.02 to C.sub.07 removed. The uncoupled resonant frequencies of each of the interior resonators C.sub.2, L.sub.2 to C.sub.7, L.sub.7 representing the resonators formed by the wafer 44 and electrodes 16 through 38 are now each detuned by the effect of the detuning capacitance C.sub.T representing the piezoelectric coupling and the interelectrode capacitances C.sub.02 to C.sub.07 which are now unshorted.

The detuning capacitance C.sub.T in each case equals C.sub.1 [C.sub.1 C.sub.0 /(C.sub.1 + C.sub.0)- C.sub.1 ]/[C.sub.1 + C.sub.1 C.sub.0 /(C.sub.1 + C.sub.0)- C.sub.1 ]. Thus C.sub.T is equal to -C.sub.1 .sup.2 /C.sub.0. Therefore, the open-circuit f.sub.oc frequency which can be tested as shown in FIG. 5 except by open circuiting the electrodes 24 and 26 and short circuiting the electrodes 12, 14, 16, 18, 28, 30, 32, 34, 36, 38, 40 and 42, includes the effects of the interelectrode capacitances C.sub.0 and the effect of the piezoelectric coupling. This constitutes a raising of the frequency from f.sub.o because the resultant capacitance is negative. The resonators formed by the electrodes 12 and 14 as well as the electrodes 40 and 42 as represented by L.sub.1, C.sub.1 and L.sub.8, C.sub.8 are alone, without the effect of C.sub.0, tuned to the frequency f.sub.oc. As before the effects of piezoelectric coupling of capacitors C.sub.01 and C.sub.08 are obviated by tuning the capacitances C.sub.01 and C.sub.08 with the inductors L.sub.i and L.sub.o to the frequency of the resonators L.sub.1, C.sub.1 and L.sub.8, C.sub.8. Here that frequency is f.sub.oc. In FIG. 6 the tuning of the uncoupled resonators, as considered alone, formed by electrodes 16 to 38, and represented by L.sub.2, C.sub.2 to L.sub.7, C.sub.7, capacitors C.sub.1, -C.sub.1 and C.sub.o, to the frequency f.sub.oc is accomplished by making the masses of the electrodes 16 to 38 to be greater than the electrodes 12, 14 and 40, 42. This reduces the uncoupled resonant frequency f.sub.o of each until the total resulting frequency of each open-circuited resonator is f.sub.oc. The increased masses achieve the previously mentioned plateback that reduces the resonant frequency of each resonator. The relative masses of each electrode pair, that is, the relative platebacks are determined not only to achieve proper tuning but to achieve the desired coupling. The greater the plateback on successive resonators, the smaller the coupling between them.

An example of the dimensions suitable for the structure of FIG. 6 is as follows. These dimensions again are only examples and should not be taken as limiting. According to this example the crystal body is composed of an AT-cut quartz crystal, 1.4 inches long, 0.400 inches wide and approximately 0.0087 inches thick. The dimensions of the electrode pairs 12 through 42 are 0.1050 inches along the long direction of the body, that is along the Z' axis, by 0.1304 inches across the Z' axis. The electrode separation d.sub.1 to d.sub.7 between the edges having the long dimensions are:

d.sub.1 = 0.0167 inches

d.sub.2 = 0.0211 inches

d.sub.3 = 0.0218 inches

d.sub.4 = 0.0220 inches

d.sub.5 = 0.0218 inches

d.sub.6 = 0.0211 inches

d.sub.7 = 0.0167 inches

The spacing dimensions have tolerances of .+-.0.0001 inches, respectively. The masses of the electrodes are such as to achieve respective platebacks of 2 percent. The resulting respective coupling coefficients k between successive pairs from left to right in FIG. 6 are 1.54.times. 10.sup.-.sup.3, 1.245.times. 10.sup.-.sup.3, 1.200.times. 10.sup.-.sup.3, 1.192.times. 10.sup.-.sup.3, 1.200.times. 10.sup.-.sup.3, 1.245.times. 10.sup.-.sup.3 and 1.54.times. 10.sup.-.sup.3. The structure of FIG. 6 passes a midband frequency of 7.5 megahertz and a passband width of about 17.5 kilohertz. The resonator inductance in each resonator is 48.5 millihenries. The filter is intended to be driven by a source impedance of 3000 ohms and when the output of the electrodes 40 and 42 is applied across a load R.sub.o of 3000 ohms. L.sub.i and L.sub.0 are 0.220 millihenries, respectively. R.sub.s = 3000.

The invention thus eliminates the limitations previously placed upon such filters by the piezoelectric coupling of the electrodes.

Because according to the invention the coupling is controlled between adjacent resonators, a wider passband is available than would be obtained otherwise with the particular crystal material.

The crystal structure of FIGS. 1, 2 and 6 is manufactured by first selecting the total bandwidth and calculating on the basis of ordinary circuit theory the needed coupling coefficients between each pair of electrodes. Electrode sizes and suitable platebacks are chosen from curves such as in FIGS. 7, 8 and 9 which have been developed for structures wherein two pairs of electrodes are coupled to each other. Where t is the thickness of the wafer and r is the width of the electrodes, r/t is generally made equal to 12 although in practice any value between 6 and 20 is usable. A value of 15t is frequently chosen as the length of the electrodes normal to the coupling axis for good suppression of other modes. The fundamental thickness shear mode frequency f is chosen from the formula f= f.sub.0 /(1- P.sub.B) where P.sub.B is the fractional shift in frequency due to mass loading of the electrodes and equal to (f- f.sub.0)/f.

The manufacture of filters such as shown in FIGS. 1, 2 and 6 starts by first cutting a wafer from a quartz crystal having the desired crystallographic orientation such as an AT cut. The wafer is then lapped and etched to a thickness t corresponding to the fundamental shear mode index frequency f. Masks with cutouts placed on each face of the crystal wafer serve for depositing the electrodes. The geometry of the electrodes is determined by considering the desired bandwidths and convenient platebacks.

The proper separation d between the electrodes of adjacent pairs may be determined from graphs such as those of FIGS. 7, 8 and 9 which show variation in coupling (f.sub.B - f.sub.A)/ f.sub.B f.sub.A for various ratios of electrode separation d to wafer thickness and for various platebacks as well as various values of r/t at one center frequency.

To obtain chosen platebacks gold or nickel is deposited such as by evaporation in layers through the masks so as to make connections possible and achieve about half the total desired plateback. Energy for measurement is applied separately to each pair of electrodes and mass added to the electrodes until a shift corresponding to the total plateback occurs. This is done until the pair resonates at the desired overall center frequency f.sub.o . During this depositing procedure the other electrode pairs are detuned by keeping them either open circuited when making the circuit of FIG. 1 or short circuited when making the circuit of FIG. 6. However, it may be necessary to obviate the effect of the other pairs by terminating them inductively or capacitively. The coupling and responses of each pair of coupled resonators are then measured as described in FIG. 5 and the desired bandwidths should prevail. Adjustments may be made by slight variation in the plateback of each pair of electrodes.

The invention furnishes a reliable energy-translating system and filter which can be constructed not only simply but to cover wide-bandwidth passbands.

The FIGS. 7, 8 and 9 are examples of empirically derived graphs for a filter having two coupled resonators operating unaffected by other resonators within a frequency range and about a center frequency. The graphs are useful for determining suitable parameters. The values f.sub.B - F.sub.A can be considered measures of the coupling k= (f.sub.B - f.sub.A)/ f.sub.B f.sub.A. The value (f.sub.B - f.sub.A)/f.sub.A is an approximation where f.sub.B is close to f.sub.A. In the graphs r is the electrode dimension in the direction along which the electrodes are aligned.

While embodiments of the invention have been shown in detail, it will be obvious to one skilled in the art that the invention may be embodied otherwise without departing from its spirit and scope.

Claims

We claim:

1. An electromechanical filter comprising, in combination,

a body of piezoelectric material,

input electrode means sandwiching a first region of said body therebetween,

output electrode means sandwiching a second region of said body therebetween,

means for applying electrical energy to said input electrode means,

means for extracting electrical energy from said output means,

intermediate electrode means sandwiching said body therebetween located between said input and output electrode means,

each of said electrode means having sufficient mass to decrease exponentially the amplitude of acoustic energy in said body as the distance from said electrode means increases,

thereby to confine said acoustic energy substantially to a limited acoustic field in said body close to said electrode means and away from the edges of said body,

each of said electrode means being spaced at a preselected distance within the acoustic field of each adjacent one of said electrode means,

the portions of said body between each adjacent pair of said electrode means being coupled acoustically as a result of the combined effect of said mass and said distance whereby energy transfer between said pairs is limited substantially to acoustic energy,

inductor means connected to said input and output electrode means, thereby to facilitate accurate shaping of the passband characteristics of said filter,

said intermediate electrode means comprising a single pair of short-circuited electrodes, said inductor means forming a respective tuning circuit with each of said input and output electrode means, and

each of said tuning circuits being tuned to the mechanical resonant frequency of a respective one of the resonators formed by said input and output means.

2. An electromechanical filter comprising, in combination,

a body of piezoelectric material,

input electrode means sandwiching a first region of said body therebetween,

output electrode means sandwiching a second region of said body therebetween,

means for applying electrical energy to said input electrode means,

means for extracting electrical energy from said output means,

intermediate electrode means sandwiching said body therebetween located between said input and output electrode means,

each of said electrode means having sufficient mass to decrease exponentially the amplitude of acoustic energy in said body as the distance from said electrode means increases,

thereby to confine said acoustic energy substantially to a limited acoustic field in said body close to said electrode means and away from the edges of said body,

each of said electrode means being spaced at a preselected distance within the acoustic field of each adjacent one of said electrode means,

the portions of said body between each adjacent pair of said electrode means being coupled acoustically as a result of the combined effect of said mass and said distance whereby energy transfer between said pairs is limited substantially to acoustic energy,

inductor means connected to said input and output electrode means, thereby to facilitate accurate shaping of the passband characteristics of said filter,

said intermediate electrode means comprising a plurality of pairs of electrodes,

said inductor means forming a respective tuning circuit with each of said input and output electrode means, and

each of said tuning circuits being tuned to the mechanical resonant frequency of a respective one of the resonators formed by said input and output means.