U.S. patent number 11,196,175 [Application Number 16/646,030] was granted by the patent office on 2021-12-07 for antenna device.
This patent grant is currently assigned to MITSUBISHI ELECTRIC CORPORATION. The grantee listed for this patent is MITSUBISHI ELECTRIC CORPORATION. Invention is credited to Yuichi Hagito, Takuma Kadoya, Yusuke Kitsukawa, Moriyasu Miyazaki, Hiroaki Sakamoto, Takashi Yanagi.
United States Patent |
11,196,175 |
Sakamoto , et al. |
December 7, 2021 |
Antenna device
Abstract
Included are: a coaxial line (10) provided so as to pass through
a second ground conductor (6), a first dielectric substrate (8),
and a second dielectric substrate (9), the coaxial line (10)
including an outer conductor (11) allowing a first ground conductor
(1), a second ground conductor (6), and a third ground conductor
(7) to be conductive thereamong; and a conductive member (15)
provided so as to pass through the first dielectric substrate (8),
the conductive member (15) allowing the first ground conductor (1)
and the second ground conductor (6) to be conductive therebetween.
An interface circuit (18) combines a plurality of signals having
mutually different phases output from each of plurality of element
antennas (3a), (3b), (3c), and (3d) and outputs the combined signal
to the coaxial line (10).
Inventors: |
Sakamoto; Hiroaki (Tokyo,
JP), Yanagi; Takashi (Tokyo, JP),
Kitsukawa; Yusuke (Tokyo, JP), Miyazaki; Moriyasu
(Tokyo, JP), Kadoya; Takuma (Tokyo, JP),
Hagito; Yuichi (Tokyo, JP) |
Applicant: |
Name |
City |
State |
Country |
Type |
MITSUBISHI ELECTRIC CORPORATION |
Tokyo |
N/A |
JP |
|
|
Assignee: |
MITSUBISHI ELECTRIC CORPORATION
(Tokyo, JP)
|
Family
ID: |
1000005981147 |
Appl.
No.: |
16/646,030 |
Filed: |
September 29, 2017 |
PCT
Filed: |
September 29, 2017 |
PCT No.: |
PCT/JP2017/035396 |
371(c)(1),(2),(4) Date: |
March 10, 2020 |
PCT
Pub. No.: |
WO2019/064470 |
PCT
Pub. Date: |
April 04, 2019 |
Prior Publication Data
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|
|
Document
Identifier |
Publication Date |
|
US 20200274251 A1 |
Aug 27, 2020 |
|
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01Q
21/24 (20130101); H01Q 1/48 (20130101); H01Q
9/42 (20130101); H01Q 21/0006 (20130101) |
Current International
Class: |
H01Q
21/00 (20060101); H01Q 9/42 (20060101); H01Q
1/48 (20060101); H01Q 21/24 (20060101) |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
|
|
|
|
|
|
|
2000-77930 |
|
Mar 2000 |
|
JP |
|
2014-135707 |
|
Jul 2014 |
|
JP |
|
Primary Examiner: Islam; Hasan
Attorney, Agent or Firm: Birch, Stewart, Kolasch &
Birch, LLP
Claims
The invention claimed is:
1. An antenna device comprising: a first ground conductor having a
first plane and a second plane; a plurality of element antennas
arranged on the first plane of the first ground conductor; a second
ground conductor arranged on the second plane of the first ground
conductor in parallel with the first ground conductor; a third
ground conductor arranged in parallel to the second ground
conductor on, of two planes of the second ground conductor, a plane
opposite to the plane on which the first ground conductor is
arranged; a first dielectric substrate arranged between the first
ground conductor and the second ground conductor; a second
dielectric substrate arranged between the second ground conductor
and the third ground conductor; a coaxial line provided so as to
pass through the second ground conductor and the first and second
dielectric substrates, the coaxial line comprising an outer
conductor allowing the first ground conductor, the second ground
conductor, and the third ground conductor to be conductive
thereamong; a conductive member provided so as to pass through the
first dielectric substrate, the conductive member allowing the
first ground conductor and the second ground conductor to be
conductive therebetween; and an interface circuit to combine a
plurality of signals having mutually different phases output from
each of the plurality of element antennas and output the combined
signal to the coaxial line, wherein each of the plurality of
element antennas has a bended point from which each of the
plurality of element antennas extends in a direction parallel to
the first ground conductor, and the direction is different among
each of the plurality of element antennas.
2. The antenna device according to claim 1, wherein the interface
circuit divides a signal transmitted by the coaxial line into a
plurality of signals having mutually different phases and outputs
each of the divided multiple signals to the plurality of element
antennas.
3. The antenna device according to claim 1, wherein the outer
conductor in the coaxial line comprises: a plurality of penetrating
members each having one end connected to the second plane of the
first ground conductor at a position surrounded by feeding points
of the plurality of element antennas; and a plurality of conductors
each inserted in one of the plurality of penetrating members, the
plurality of conductors allowing the first ground conductor, the
second ground conductor, and the third ground conductor to be
conductive thereamong, and the coaxial line comprises the outer
conductor and an inner conductor arranged at a position surrounded
by the plurality of penetrating members.
4. The antenna device according to claim 1, wherein the first to
third ground conductors are flat plates having square planar
shapes, a length of each side of the second ground conductor is
equal to a length of half a wavelength at a resonance frequency of
the plurality of element antennas, and a length of each side of the
third ground conductor is longer than or equal to the length of
half the wavelength at the resonance frequency of the plurality of
element antennas.
5. The antenna device according to claim 1, wherein the first
ground conductor is a flat plate having a square planar shape, four
element antennas are arranged on the first plane of the first
ground conductor as the plurality of element antennas, each of the
four element antennas is an inverted L antenna having a bended
point between a feeding point and a tip, and in the four element
antennas, each of tip portions extending from the bended points to
the tips is parallel to the first plane of the first ground
conductor, and directions from the bended points to the tips are
different from each other by 90 degrees and are parallel to any one
of sides of the first ground conductor.
6. The antenna device according to claim 1, wherein the first
ground conductor is a flat plate having a square planar shape, four
element antennas are arranged on the first plane of the first
ground conductor as the plurality of element antennas, each of the
four element antennas is an inverted F antenna having a feeding
point and a connection point with the first plane of the first
ground conductor, and in the four element antennas, each of tip
portions extending from bended points between the feeding points
and tips to the tips is parallel to the first plane of the first
ground conductor, and directions from the bended points to the tips
are different from each other by 90 degrees and are parallel to any
one of sides of the first ground conductor.
7. The antenna device according to claim 1, wherein the first
ground conductor is a flat plate having a square planar shape, four
element antennas are arranged on the first plane of the first
ground conductor as the plurality of element antennas, each of the
four element antennas is a folded monopole antenna having a feeding
point and a connection point with the first plane of the first
ground conductor, and in the four element antennas, each of
portions extending from bended points between the feeding points
and folded points to the folded points is parallel to the first
plane of the first ground conductor, and directions from the bended
points to the folded points are different from each other by 90
degrees and are parallel to any one of sides of the first ground
conductor.
8. The antenna device according to claim 1, wherein a passive
element corresponding to each of the plurality of element antennas
is arranged on the first plane of the first ground conductor.
9. The antenna device according to claim 1, wherein each of four
sides of the second ground conductor having a square planar shape
is notched.
10. The antenna device according to claim 1, further comprising: a
third dielectric substrate arranged on the first plane of the first
ground conductor, wherein the plurality of element antennas is
formed in the third dielectric substrate.
11. The antenna device according to claim 1, comprising: a fourth
ground conductor arranged in parallel with the third ground
conductor on, of two planes of the third ground conductor, a plane
opposite to the plane on the side on which the second ground
conductor is arranged; a communication component circuit attached
to, of two planes of the fourth ground conductor, a plane opposite
to the plane on which the third ground conductor is arranged; and a
first metal housing shielding the communication component circuit
from surroundings thereof.
12. The antenna device according to claim 11, further comprising: a
second metal housing arranged so as to surround the first metal
housing, wherein a space between the first metal housing and the
second metal housing is filled with a resin member.
Description
TECHNICAL FIELD
The present invention relates to an antenna device including a
plurality of element antennas.
BACKGROUND ART
Some of terminals for receiving polarized waves transmitted from
satellite telephone service or global positioning system (GPS)
satellites use circularly polarized wave antennas in order to avoid
a polarized wave loss from growing even when a terminal user
moves.
Examples of circularly polarized wave antennas include spiral
antennas and patch antennas. However, it is known that a circularly
polarized wave antenna such as a spiral antenna is increased in
size if an attempt is made to broaden the bandwidth of the
antenna.
Moreover, for example when a polarized wave transmitted from a GPS
satellite is reflected by the ground or a building, the polarized
wave may be changed to reverse rotation.
In a case where the polarized wave transmitted from the GPS
satellite is a right-handed circularly polarized wave (RHCP), the
RHCP may change to a left-handed circularly polarized wave
(LHCP).
It is known that when a circularly polarized wave antenna such as a
spiral antenna is reduced in size, a back lobe, which is a cross
polarized wave extending backward from the antenna, increases. In a
case where the polarized wave transmitted from the GPS satellite is
RHCP, a back lobe which is a cross polarized wave is LHCP.
For this reason, reducing the size of a circularly polarized wave
antenna increases the possibility for the circularly polarized wave
antenna to receive unwanted LHCPs, which may deteriorate the
positioning performance based on polarized waves transmitted from
the GPS satellites.
Since reducing the size of a circularly polarized wave antenna
increases the possibility of receiving unnecessary back lobes, a
large-sized circularly polarized wave antenna is generally used. In
a case where it is highly desired to reducing the size of a
circularly polarized wave antenna, however, reception of
unnecessary back lobes may be suppressed by providing a large
ground plate separately.
However, in a case where a large ground plate is provided
separately, the entire antenna device including the circularly
polarized wave antenna becomes larger.
Patent Literature 1 below discloses an antenna device in which
reception of unnecessary back lobes is suppressed without
separately providing a large ground plate.
The antenna device disclosed in Patent Literature 1 suppresses
reception of unnecessary back lobes by providing a choke structure
on the bottom surface of a radiation conductor.
The choke structure provided on the bottom surface of the radiation
conductor has two conductor plates arranged in parallel, and the
center portions of the two conductor plates are thicker than the
edges of the two conductor plates.
By allowing the center portions of the two conductor plates and the
edges of the two conductor plates to have different thicknesses, it
becomes possible to adjust the electrical length of the choke
structure depending on the frequency of an unnecessary back
lobe.
CITATION LIST
Patent Literature
Patent Literature 1: JP 2014-135707 A
SUMMARY OF INVENTION
Technical Problem
Since a conventional antenna device is configured as described
above, reception of unnecessary back lobes can be suppressed
without providing a large ground plate separately.
However, since the choke structure provided instead of a large
ground plate has a complicated structure in which the center
portions of the two conductor plates and the edges of the two
conductor plates have different thicknesses, producing the antenna
devices is disadvantageously troublesome.
The present invention has been devised to solve the disadvantage as
described above, and an object of the invention is to obtain an
antenna device capable of adjusting the resonance frequency and
suppressing reception of unnecessary back lobes without mounting a
complicated choke structure.
Solution to Problem
An antenna device according to the present invention includes: a
first ground conductor having a first plane and a second plane; a
plurality of element antennas arranged on the first plane of the
first ground conductor; a second ground conductor arranged on the
second plane of the first ground conductor in parallel with the
first ground conductor; a third ground conductor arranged in
parallel to the second ground conductor on, of two planes of the
second ground conductor, a plane opposite to the plane on which the
first ground conductor is arranged; a first dielectric substrate
arranged between the first ground conductor and the second ground
conductor; a second dielectric substrate arranged between the
second ground conductor and the third ground conductor; a coaxial
line provided so as to pass through the second ground conductor and
the first and second dielectric substrates, the coaxial line
including an outer conductor allowing the first ground conductor,
the second ground conductor, and the third ground conductor to be
conductive thereamong; a conductive member provided so as to pass
through the first dielectric substrate, the conductive member
allowing the first ground conductor and the second ground conductor
to be conductive therebetween; and an interface circuit for
combining a plurality of signals having mutually different phases
output from each of the plurality of element antennas and
outputting the combined signal to the coaxial line.
Advantageous Effects of Invention
According to this invention, included are: a coaxial line provided
so as to pass through a second ground conductor and first and
second dielectric substrates, the coaxial line including an outer
conductor allowing the first ground conductor, the second ground
conductor, and the third ground conductor to be conductive
thereamong; and a conductive member provided so as to pass through
the first dielectric substrate, the conductive member allowing the
first ground conductor and the second ground conductor to be
conductive therebetween, and an interface circuit combines a
plurality of signals having mutually different phases output from
each of the plurality of element antennas and outputs the combined
signal to the coaxial line. This allows the resonance frequency to
be adjusted to suppress reception of unwanted back lobes without
mounting a complicated choke structure.
BRIEF DESCRIPTION OF DRAWINGS
FIG. 1 is a perspective view illustrating an antenna device
according to a first embodiment of the present invention.
FIG. 2 is a cross-sectional side view of the antenna device of FIG.
1 as viewed from direction A.
FIG. 3 is a plan view illustrating feeding points 4a, 4b, 4c, and
4d of element antennas 3a, 3b, 3c, and 3d in a first plane 1a of a
first ground conductor 1, a coaxial line 10, and an interface
circuit 18 on a first plane 1a of a first ground conductor 1.
FIG. 4 is a perspective view illustrating an antenna device not
including a third ground conductor 7 nor a second dielectric
substrate 9.
FIG. 5 is a cross-sectional side view of the antenna device of FIG.
4 as viewed from direction A.
FIG. 6 is an explanatory graph illustrating the RHCP gain and the
LHCP gain in an antenna device in which a first dielectric
substrate 8, a first ground conductor 1, and a second ground
conductor 6 each have a short side.
FIG. 7 is an explanatory diagram illustrating a simple model
including current sources (J1 to J4) and magnetic current sources
(M1 to M4).
FIG. 8 is an explanatory graph illustrating a simulation result of
a correspondence relationship between the phase difference
.DELTA..phi. and peak values of radiation patterns.
FIG. 9 is an explanatory diagram illustrating the RHCP gain and the
LHCP gain in an antenna device.
FIG. 10A is an explanatory diagram illustrating an example in which
element antennas are inverted F antennas.
FIG. 10B is an explanatory diagram illustrating an example in which
element antennas are folded monopole antennas.
FIG. 11A is an explanatory diagram illustrating an example in which
element antennas that are inverted L antennas and a passive element
30 are included.
FIG. 11B is an explanatory diagram illustrating an example in which
element antennas that are inverted F antennas and a passive element
30 are included.
FIG. 11C is an explanatory diagram illustrating an example in which
element antennas that are folded monopole antennas and a passive
element 30 are included.
FIG. 12 is a plan view illustrating a first ground conductor 1 and
a first dielectric substrate 8 having circular planar shapes.
FIG. 13 is a cross-sectional side view illustrating another antenna
device according to the first embodiment of the invention.
FIG. 14 is a plan view illustrating a planar shape of a second
ground conductor 6 in the antenna device according to a second
embodiment of the present invention.
FIG. 15 is a cross-sectional side view illustrating an antenna
device according to a third embodiment of the invention.
FIG. 16 is a plan view illustrating a top surface of the antenna
device according to the third embodiment of the present
invention.
FIG. 17 is a cross-sectional side view illustrating an antenna
device according to a fourth embodiment of the invention.
FIG. 18 is a cross-sectional side view illustrating an antenna
device according to a fifth embodiment of the invention.
DESCRIPTION OF EMBODIMENTS
To describe the present invention further in detail, embodiments
for carrying out the present invention will be described below with
reference to the accompanying drawings.
First Embodiment
FIG. 1 is a perspective view illustrating an antenna device
according to a first embodiment of the present invention.
FIG. 2 is a cross-sectional side view of the antenna device of FIG.
1 as viewed from direction A.
FIG. 3 is a plan view illustrating feeding points 4a, 4b, 4c, and
4d of element antennas 3a, 3b, 3c, and 3d, a coaxial line 10, and
an interface circuit 18 on a first plane 1a of a first ground
conductor 1.
In FIGS. 1 to 3, the first ground conductor 1 has the first plane
1a and a second plane 1b.
The first ground conductor 1 is a flat plate having a square planar
shape.
A circularly polarized wave transmitting/receiving unit 2 is
arranged on the first plane 1a of the first ground conductor 1.
The circularly polarized wave transmitting/receiving unit 2
includes the element antennas 3a, 3b, 3c, and 3d that can transmit
and receive circularly polarized waves.
In this first embodiment, an example will be explained in which the
circularly polarized wave transmitting/receiving unit 2 has four
element antennas 3a, 3b, 3c, and 3d as element antennas; however,
the number of element antennas is only required to be plural, and
is not limited to four.
The feeding points 4a, 4b, 4c, and 4d of the element antennas 3a,
3b, 3c, and 3d indicate, for example, positions where a signal
output from the interface circuit 18 is input when a circularly
polarized wave is transmitted. Although the feeding points 4a, 4b,
4c, and 4d are drawn in FIGS. 1 to 3; however, the feeding points
4a, 4b, 4c, and 4d are not formed as physical components of the
antenna device.
The element antennas 3a, 3b, 3c, and 3d are inverted L antennas
having bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b
between the feeding points 4a, 4b, 4c, and 4d and tips 5a, 5b, 5c,
and 5d, respectively.
The total length of each of the element antennas 3a, 3b, 3c, and 3d
is about a quarter wavelength at the resonance frequency.
In the element antennas 3a, 3b, 3c, and 3d, each of the tip
portions extending from the bended points 3a.sub.b, 3b.sub.b,
3c.sub.b, and 3d.sub.b to the tips 5a, 5b, 5c, and 5d is parallel
to the first plane 1a of the first ground conductor 1.
Moreover, in the element antennas 3a, 3b, 3c, and 3d, directions
from the bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b
to the tips 5a, 5b, 5c, and 5d are different from each other by 90
degrees, and are parallel to one of the sides of the first ground
conductor 1.
In FIG. 1, the direction from the bended point 3a.sub.b to the tip
5a is parallel to the lower side of the first ground conductor 1 on
the paper, and the direction from the bended point 3b.sub.b to the
tip 5b is parallel to the left side of the first ground conductor 1
on the paper.
The direction from the bended point 3c.sub.b to the tip 5c is
parallel to the upper side of the first ground conductor 1 on the
paper, and the direction from the bended point 3d.sub.b to the tip
5d is parallel to the right side of the first ground conductor 1 on
the paper.
A second ground conductor 6 is arranged in parallel with the first
ground conductor 1 on a second plane 1b side of the first ground
conductor 1.
The second ground conductor 6 is a flat plate having a square
planar shape, and the length of each side of the second ground
conductor 6 is half a wavelength at the resonance frequency of the
element antennas 3a, 3b, 3c, and 3d.
Note that the length of each side of the second ground conductor 6
may be the length that completely matches the length of the half
wavelength at the resonance frequency and may also be a length that
substantially matches the length of the half wavelength at the
resonance frequency.
A third ground conductor 7 is arranged in parallel with the second
ground conductor 6 on the plane opposite to the plane on which the
first ground conductor 1 is arranged out of the two planes of the
second ground conductor 6.
The third ground conductor 7 is a flat plate having a square planar
shape, and the length of each side of the third ground conductor 7
is longer than or equal to half a wavelength at the resonance
frequency of the element antennas 3a, 3b, 3c, and 3d.
A first dielectric substrate 8 is arranged between the first ground
conductor 1 and the second ground conductor 6.
A second dielectric substrate 9 is arranged between the second
ground conductor 6 and the third ground conductor 7.
The length of each side of the second dielectric substrate 9 is
longer than or equal to the length of each side of the third ground
conductor 7 since the second ground conductor 6 and the third
ground conductor 7 are copper foil patterns on the second
dielectric substrate 9.
The coaxial line 10 includes an outer conductor 11 and an inner
conductor 14. Although the coaxial line 10 is illustrated in FIGS.
2 and 3, the coaxial line 10 is not illustrated in FIG. 1 to
simplify the drawing.
The outer conductor 11 is provided so as to pass through the second
ground conductor 6, the first dielectric substrate 8, and the
second dielectric substrate 9, and allows the first ground
conductor 1, the second ground conductor 6, and the third ground
conductor 7 to be conductive thereamong.
The outer conductor 11 includes a penetrating member 12 and a
conductor 13, and one end thereof is connected to the second plane
1b of the first ground conductor 1 at a position surrounded by the
feeding points 4a, 4b, 4c, and 4d of the element antennas 3a, 3b,
3c, and 3d.
In FIG. 3, an example is illustrated in which seven outer
conductors 11 are arranged.
The penetrating member 12 is a through-hole member arranged at a
position surrounded by the feeding points 4a, 4b, 4c, and 4d in the
element antennas 3a, 3b, 3c, and 3d on the second plane 1b of the
first ground conductor 1.
The conductor 13 is a metal member that is inserted in the
penetrating member 12 to allow the first ground conductor 1, the
second ground conductor 6, and the third ground conductor 7 to be
conductive thereamong.
The inner conductor 14 is arranged at a position surrounded by the
plurality of outer conductors 11, and one end 14a of the inner
conductor 14 is connected to a 180-degree hybrid 19 of the
interface circuit 18.
The other end 14b of the inner conductor 14 is connected to a
circuit (not illustrated) for inputting and outputting signals.
A conductive member 15 includes a penetrating member 16 and a
conductor 17, and one end thereof is connected to the second plane
1b of the first ground conductor 1 at a position surrounding the
feeding points 4a, 4b, 4c, and 4d of the element antennas 3a, 3b,
3c, and 3d.
Although, in FIG. 2, an example is illustrated in which two
conductive members 15 are arranged, tens or hundreds of conductive
members 15 are often arranged in practice.
The conductive members 15 are provided so as to pass through the
first dielectric substrate 8 and allow the first ground conductor 1
and the second ground conductor 6 to be conductive
therebetween.
The penetrating members 16 are through-hole members arranged at
positions surrounding the feeding points 4a, 4b, 4c, and 4d in the
element antennas 3a, 3b, 3c, and 3d on the second plane 1b of the
first ground conductor 1.
The conductor 17 is a metal member that is inserted in a
penetrating member 16 to allow the first ground conductor 1 and the
second ground conductor 6 to be conductive therebetween.
The interface circuit 18 includes the 180-degree hybrid 19 and
90-degree hybrids 20 and 21, and is patterned on the first plane 1a
of the first ground conductor 1 by etching.
The interface circuit 18 combines four signals having mutually
different phases output from each of the feeding points 4a, 4b, 4c,
and 4d of the element antennas 3a, 3b, 3c, and 3d and outputs the
combined signal to the coaxial line 10 when the element antennas
3a, 3b, 3c, and 3d are used as reception antennas.
The interface circuit 18 divides a signal transmitted by the
coaxial line 10 into four signals having mutually different phases
and outputs each of the divided four signals to one of the feeding
points 4a, 4b, 4c, and 4d of the element antennas 3a, 3b, 3c, and
3d when the element antennas 3a, 3b, 3c, and 3d are used as
transmission antennas.
Although the interface circuit 18 is illustrated in FIG. 3, the
interface circuit 18 is not illustrated in FIG. 1 nor 2 to simplify
the drawings.
The 180-degree hybrid 19 combines, for example, a signal having a
phase of 0 degrees output from the 90-degree hybrid 20 and, for
example, a signal having a phase of 180 degrees output from the
90-degree hybrid 21 and outputs the combined signal to the coaxial
line 10 when the element antennas 3a, 3b, 3c, and 3d are used as
reception antennas.
The 180-degree hybrid 19 divides a signal transmitted by the
coaxial line 10 into two signals having phases that are 180 degrees
different from each other, and outputs one of the divided signals
to the 90-degree hybrid 20 and the other divided signal to the
90-degree hybrid 21 when the element antennas 3a, 3b, 3c, and 3d
are used as transmission antennas.
For example, in a case where the phase of one of the divided
signals is 0 degrees, the phase of the signal output from the
180-degree hybrid 19 to the 90-degree hybrid 20 is 0 degrees, and
the phase of the signal output from the 180-degree hybrid 19 to the
90-degree hybrid 21 is 180 degrees.
The 90-degree hybrid 20 combines, for example, a signal having a
phase of 0 degrees output from the feeding point 4a of the element
antenna 3a and, for example, a signal having a phase of 90 degrees
output from the feeding point 4b of the element antenna 3b and
outputs a synthesized signal having a phase of 0 degrees to the
180-degree hybrid 19 when the element antennas 3a, 3b, 3c, and 3d
are used as reception antennas.
The 90-degree hybrid 20 divides, for example, a signal having a
phase of 0 degrees output from the 180-degree hybrid 19 into a
signal having a phase of 0 degrees and a signal having a phase of
90 degrees, and outputs the divided signal having the phase of 0
degrees to the feeding point 4a of the element antenna 3a and the
divided signal having the phase of 90 degrees to the feeding point
4b of the element antenna 3b when the element antennas 3a, 3b, 3c,
and 3d are used as transmission antennas.
The 90-degree hybrid 21 combines a signal having, for example, a
phase of 180 degrees output from the feeding point 4c of the
element antenna 3c and a signal having, for example, a phase of 270
degrees output from the feeding point 4d of the element antenna 3d
and outputs a synthesized signal having a phase of 180 degrees to
the 180-degree hybrid 19 when the element antennas 3a, 3b, 3c, and
3d are used as reception antennas.
The 90-degree hybrid 21 divides, for example, a signal having a
phase of 180 degrees output from the 180-degree hybrid 19 into a
signal having a phase of 180 degrees and a signal having a phase of
270 degrees, and outputs the divided signal having the phase of 180
degrees to the feeding point 4c of the element antenna 3c and the
divided signal having the phase of 270 degrees to the feeding point
4d of the element antenna 3d when the element antennas 3a, 3b, 3c,
and 3d are used as transmission antennas.
In the first embodiment, the portion sandwiched between the second
ground conductor 6 and the third ground conductor 7 operates as a
microstrip resonator 22.
Next, the operation will be described.
Since the operation when the element antennas 3a, 3b, 3c, and 3d
are used as transmission antennas and the operation when the
element antennas 3a, 3b, 3c, and 3d are used as reception antennas
are reversible, here, the operation when the element antennas 3a,
3b, 3c, and 3d are used as transmission antennas will be described
representatively.
When a signal is given from the circuit not illustrated to the
other end 14b of the inner conductor 14 in the coaxial line 10, the
signal given from the circuit not illustrated is transmitted to the
one end 14a of the coaxial line 10 and then is transmitted to the
interface circuit 18.
Here, for convenience of explanation, it is assumed that the phase
of the signal output from the one end 14a of the coaxial line 10 to
the interface circuit 18 is 0 degrees.
The 180-degree hybrid 19 of the interface circuit 18 divides the
signal having the phase of 0 degrees output from the one end 14a of
the coaxial line 10 into two signals having phases that are 180
degrees different from each other, and outputs a signal having a
phase of 0 degrees to the 90-degree hybrid 20 and a signal having a
phase of 180 degrees to the 90-degree hybrid 21.
The 90-degree hybrid 20 divides the signal having the phase of 0
degrees output from the 180-degree hybrid 19 into two signals
having phases that are 90 degrees different from each other, and
outputs a signal having a phase of 0 degrees to the feeding point
4a of the element antenna 3a and a signal having a phase of 90
degrees to the feeding point 4b of the element antenna 3b.
The 90-degree hybrid 21 divides the signal having the phase of 180
degrees output from the 180-degree hybrid 19 into two signals
having phases that are 90 degrees different from each other, and
outputs a signal having a phase of 180 degrees to the feeding point
4c of the element antenna 3c and a signal having a phase of 270
degrees to the feeding point 4d of the element antenna 3d.
As a result, the element antennas 3a, 3b, 3c, and 3d of the
circularly polarized wave transmitting/receiving unit 2 are
provided with signals whose phases are different from each other by
90 degrees, and an electromagnetic wave corresponding to the
signals is radiated into a space as a result of the resonance
phenomenon generated when the signals are transmitted through the
element antennas 3a, 3b, 3c, and 3d.
Since the phases of the signals transmitted through the element
antennas 3a, 3b, 3c, and 3d are different from each other by 90
degrees, RHCP which is a desired electromagnetic wave is radiated
in the zenith direction (0 deg) illustrated in FIG. 2 and LHCP
which is an unwanted electromagnetic wave is radiated in the ground
direction (.+-.90 deg).
The antenna device includes the third ground conductor 7 and the
second dielectric substrate 9 in the first embodiment; however, a
case is assumed as illustrated in FIGS. 4 and 5 in which the
antenna device does not include the third ground conductor 7 nor
the second dielectric substrate 9.
FIG. 4 is a perspective view illustrating an antenna device not
including the third ground conductor 7 nor the second dielectric
substrate 9.
FIG. 5 is a cross-sectional side view of the antenna device of FIG.
4 as viewed from direction A.
In a case of an antenna device in which the length of each side of
the first dielectric substrate 8, the first ground conductor 1, and
the second ground conductor 6 is short, the values of the RHCP gain
and the LHCP gain are substantially the same as illustrated in FIG.
6. As an example in which the length of each side of the second
ground conductor 6 is short, the length of half a wavelength at the
resonance frequency of the element antennas 3a, 3b, 3c, and 3d is
conceivable.
FIG. 6 is an explanatory graph illustrating the RHCP gain and the
LHCP gain in an antenna device in which each side of the first
dielectric substrate 8, the first ground conductor 1, and the
second ground conductor 6 is short.
The horizontal axis in FIG. 6 represents the zenith angles of RHCP
and LHCP, and the vertical axis in FIG. 6 represents the gains of
the RHCP and the LHCP.
For example, when an RHCP signal is transmitted to the ground from
a GPS satellite or a quasi-zenith satellite, the RHCP signal is
reflected by the ground, a building, etc., and the RHCP signal is
inverted to generate LHCP.
Since the antenna device, in which each side of the first
dielectric substrate 8, the first ground conductor 1, and the
second ground conductor 6 is short, has substantially the same
value for the RHCP gain and the LHCP gain, there is a high
possibility that an LHCP signal, which is an unwanted wave, is
received erroneously when the antenna device is used for receiving
an RHCP signal transmitted from the GPS satellite or the
quasi-zenith satellite to the ground. Therefore, the possibility of
incurring degradation of the positioning performance based on the
RHCP increases.
In the first embodiment, the antenna device includes the third
ground conductor 7 and the second dielectric substrate 9 in order
to reduce the possibility of erroneously receiving an LHCP signal
that is an unwanted wave even in the case where the length of each
side of the first dielectric substrate 8, the first ground
conductor 1, and the second ground conductor 6 is small.
In the first embodiment, the length of each side of the second
ground conductor 6 equals the length of half a wavelength at the
resonance frequency of the element antennas 3a, 3b, 3c, and 3d.
The length of each side of the third ground conductor 7 is longer
than or equal to the length of the half wavelength at the resonance
frequency of the element antennas 3a, 3b, 3c, and 3d.
Moreover, the length of each side of the second dielectric
substrate 9 is longer than or equal to the length of each side of
the third ground conductor 7.
Therefore, a resonance phenomenon occurs in the microstrip
resonator 22 due to electromagnetic waves transmitted and received
by the element antennas 3a, 3b, 3c, and 3d.
Therefore, by adjusting the resonance frequency of the element
antennas 3a, 3b, 3c, and 3d and the resonance frequency of the
microstrip resonator 22, a broadband impedance characteristic can
be obtained. Moreover, not only that a broadband impedance
characteristic can be obtained, but also the broadband impedance
characteristic can be maintained even when the antenna device is
installed on a large ground plate.
That is, although the resonance frequency of the microstrip
resonator 22 slightly changes being affected by the fringing effect
when the antenna device is installed on a large ground plate, but
there is no significant difference from the case where the antenna
device is not installed on a large ground plate. Therefore, even
when the antenna device is installed on a large ground plate, a
broadband impedance characteristic can be maintained.
Note that the wider the gap between the second ground conductor 6
and the third ground conductor 7 is, the wider the band of the
microstrip resonator 22 becomes, and thus a broadband impedance
characteristic can be obtained.
Furthermore, when the resonance frequency of the element antennas
3a, 3b, 3c, and 3d and the resonance frequency of the microstrip
resonator 22 are adjusted to the same level, the radiation pattern
of the antenna device is obtained by superimposing the radiation
pattern of the circularly polarized wave transmitting/receiving
unit 2 as a current source and the radiation pattern of the
microstrip resonator 22 as a magnetic current source.
As illustrated in FIG. 7, the radiation pattern obtained by the
antenna device can be expressed by a simple model including current
sources (J1 to J4) and magnetic current sources (M1 to M4).
FIG. 7 is an explanatory diagram illustrating a simple model
including the current sources (J1 to J4) and the magnetic current
sources (M1 to M4).
In this example, the phase differences between the current sources
(J1 to J4) are 90 degrees each and the phase differences between
the magnetic current sources (M1 to M4) are 90 degrees each so that
RHCP is radiated in the zenith direction.
It is also assumed that the amplitudes of the current sources (J1
to J4) and the amplitudes of the magnetic current sources (M1 to
M4) are all equal and that a phase difference between a current
source (Jn: n=1, 2, 3, 4) and a magnetic current source (Mn: n=1,
2, 3, 4) is .DELTA..phi..
Although the positions of the current sources and the magnetic
current sources appear to be different in FIG. 7, they are assumed
to be in the same position.
By performing electromagnetic field analysis on the basis of the
relationship illustrated in FIG. 7, the relationship between the
phase difference .DELTA..phi. between a current source (Jn) and a
magnetic current source (Mn) and the peak value of the radiation
pattern is simulated.
FIG. 8 is an explanatory graph illustrating a simulation result of
the correspondence relationship between the phase difference
.DELTA..phi. and the peak values of the radiation patterns.
The horizontal axis in FIG. 8 represents the phase difference
.DELTA..phi. between the current source (Jn) and the magnetic
current source (Mn), and the vertical axis in FIG. 8 represents the
peak value of the radiation patterns.
FIG. 8 indicates that LHCP can be suppressed when the phase
difference .DELTA..phi. is positive and the phase of the magnetic
current source (Mn) is delayed from the phase of the current source
(Jn). When .DELTA..phi.=90 degrees, LHCP is most suppressed.
The relationship between the phase difference .DELTA..phi. and the
peak values of the radiation patterns is dependent on the physical
positions of the element antennas 3a, 3b, 3c, and 3d but also
contributes to the phase centers of the element antennas 3a, 3b,
3c, and 3d. Therefore, it becomes possible to adjust the
suppression amount of LHCP by adopting inverted L antennas as the
element antennas 3a, 3b, 3c, and 3d to allow the phase centers of
the element antennas 3a, 3b, 3c, and 3d to move in the vertical
direction that is the zenith direction (0 deg).
Specifically, the suppression amount of LHCP can be adjusted by
changing the shapes of the element antennas 3a, 3b, 3c, and 3d. As
a result, as illustrated in FIG. 9, a radiation pattern having a
high gain and a low cross-polarization (LHCP) can be obtained.
FIG. 9 is an explanatory diagram illustrating the RHCP gain and the
LHCP gain in the antenna device.
The horizontal axis in FIG. 9 represents the zenith angle of RHCP
and LHCP, and the vertical axis in FIG. 9 represents the RHCP and
LHCP gains.
In FIG. 9, the phase difference .DELTA..phi. is adjusted to 90
degrees, and the LHCP is most suppressed at a phase of 0
degrees.
As is apparent from the above, according to the first embodiment,
included are: the coaxial line 10 provided so as to pass through
the second ground conductor 6 and the first and second dielectric
substrates 8 and 9, the coaxial line 10 including the outer
conductor 11 allowing the first ground conductor 1, the second
ground conductor 6, and the third ground conductor 7 to be
conductive thereamong; and the conductive member 15 provided so as
to pass through the first dielectric substrate 8, the conductive
member 15 allowing the first ground conductor 1 and the second
ground conductor 6 to be conductive therebetween, and the interface
circuit 18 combines a plurality of signals having mutually
different phases output from each of the plurality of element
antennas 3a, 3b, 3c, and 3d and outputs the combined signal to the
coaxial line 10. This allows the resonance frequency to be adjusted
to suppress reception of unnecessary back lobes without mounting a
complex choke structure.
Although the example in which the element antennas 3a, 3b, 3c, and
3d are inverted L antennas is described in the first embodiment,
the antennas are only required to have element shapes having
directivity in the zenith direction, and the element antennas 3a,
3b, 3c, and 3d are not limited to inverted L antennas.
For example, the element antennas 3a, 3b, 3c, and 3d may be
inverted antennas as illustrated in FIG. 10A, or may be folded
monopole antennas as illustrated in FIG. 10B.
FIG. 10A is an explanatory diagram illustrating an example in which
the element antennas are inverted F antennas, and FIG. 10B is an
explanatory diagram illustrating an example in which the element
antennas are folded monopole antennas.
Like the inverted L antennas, the inverted antennas have feeding
points 4a, 4b, 4c, and 4d, and also have connection points with the
first plane 1a of the first ground conductor 1.
In the case where the element antennas 3a, 3b, 3c, and 3d are
inverted antennas, the lengths from the feeding points 4a, 4b, 4c,
and 4d to the tips 5a, 5b, 5c, and 5d are about a quarter
wavelength at the resonance frequency.
In the inverted antennas, each of the tip portions extending from
bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b to tips
5a, 5b, 5c, and 5d is parallel to the first plane 1a of the first
ground conductor 1.
Moreover, in the inverted F typo antennas, directions from the
bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b to the
tips 5a, 5b, 5c, and 5d are different from each other by 90
degrees, and are parallel to one of the sides of the first ground
conductor 1.
Like the inverted L antennas, the folded monopole antennas have
feeding points 4a, 4b, 4c, and 4d, and also have connection points
with the first plane 1a of the first ground conductor 1.
In the case where the element antennas 3a, 3b, 3c, and 3d are
folded monopole antennas, the lengths from the feeding points 4a,
4b, 4c, and 4d to the connection points are about half a wavelength
at the resonance frequency.
In the folded monopole antennas, each of the portions extending
from bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b to
folded points is parallel to the first plane 1a of the first ground
conductor 1.
Moreover, in the folded monopole antennas, directions from the
bended points 3a.sub.b, 3b.sub.b, 3c.sub.b, and 3d.sub.b to the
folded points are different from each other by 90 degrees, and are
parallel to any one of the sides of the first ground conductor
1.
Furthermore, since the element antennas 3a, 3b, 3c, and 3d are only
required to be element-type antennas having directivity in the
zenith direction, antennas such as loop antennas, helical antennas
or meander antennas may be used.
In the first embodiment, the antenna device having four feeding
points is illustrated; however, for example, an antenna device
having two feeding points or one feeding point may be used.
In the first embodiment, the example in which the circularly
polarized wave transmitting/receiving unit 2 includes the element
antennas 3a, 3b, 3c, and 3d is illustrated; however, a passive
element 30 corresponding to each of the element antennas 3a, 3b,
3c, and 3d may be included as illustrated in FIG. 11.
FIG. 11A is an explanatory diagram illustrating an example in which
element antennas that are inverted L antennas and passive elements
30 are included, FIG. 11B is an explanatory diagram illustrating an
example in which element antennas that are inverted F antennas and
passive elements 30 are included, and FIG. 11C is an explanatory
diagram illustrating an example in which element antennas that are
folded monopole antennas and passive elements 30 are included.
With the circularly polarized wave transmitting/receiving unit 2
including the passive elements 30 in addition to the element
antennas 3a, 3b, 3c, and 3d, the antenna device functions as a
multiband antenna that resonates in a plurality of bands.
In the case of the multiband antenna using the passive elements 30,
it is possible to adjust the coupling amount of the element
antennas 3a, 3b, 3c, and 3d. For this reason, for example, it is
possible to suppress unwanted waves of long term evolution (LTE) of
the 1.5 GHz band that is between multiple frequencies used in the
quasi-zenith satellites.
In the case of using the passive elements 30, there is an advantage
that the cost can be reduced as compared with a case where a
high-performance filter is used to suppress unwanted waves of the
LTE.
Although the example is illustrated in the first embodiment in
which a signal is given from the other end 14b of the inner
conductor 14 in the coaxial line 10 when the element antennas 3a,
3b, 3c, and 3d are used as transmission antennas, a signal may be
given from a side surface of the first ground conductor 1, for
example.
In FIG. 2, the side surface of the first ground conductor 1 may be,
for example, on the left side or the right side, on the paper, of
the first ground conductor 1.
In the case where a signal is given from the side surface of the
first ground conductor 1, the coaxial line 10 penetrating through
the substrates becomes unnecessary. However, this results in
asymmetry in the structure, thus deteriorating the axial ratio, and
thus it is desirable that a signal is given from the other end 14b
of the inner conductor 14 in the coaxial line 10.
In the first embodiment, the example is illustrated in which the
interface circuit 18 is patterned on the first plane 1a of the
first ground conductor 1 by etching.
However, this is merely an example. For example, the interface
circuit 18 may be formed using a chip component or the like.
In the first embodiment, the example is illustrated in which the
coaxial line 10 capable of signal transmission is formed with the
plurality of outer conductors 11 arranged at positions surrounding
the inner conductor 14.
In this case, although it is desirable that the intervals between
the multiple outer conductors 11 be dense, if the intervals are too
narrow, a line drawn from the inner conductor 14 in the coaxial
line 10 to the interface circuit 18 cannot be formed.
For this reason, the plurality of outer conductors 11 is arranged
in a C shape in the first embodiment as illustrated in FIG. 3.
Specifically, an interval between two outer conductors 11 is
widened only at a position where a line is drawn from the inner
conductor 14 in the coaxial line 10 to the interface circuit 18 as
compared to other positions.
Although the example is illustrated in the first embodiment in
which the planar shapes of the first ground conductor 1, the second
ground conductor 6, the third ground conductor 7, the first
dielectric substrate 8, and the second dielectric substrate 9 is a
square, the present embodiment is not limited to the example of
square planar shapes. For example, as illustrated in FIG. 12, the
planar shapes of the first ground conductor 1, the second ground
conductor 6, the third ground conductor 7, the first dielectric
substrate 8, and the second dielectric substrate 9 may be
circular.
FIG. 12 is a plan view illustrating a first ground conductor 1 and
a first dielectric substrate 8 having circular planar shapes.
In FIG. 12, for simplification of the drawing, feeding points 4a,
4b, 4c, and 4d of element antennas 3a, 3b, 3c, and 3d and an
interface circuit 18 are not illustrated.
The example is illustrated in the first embodiment in which the
first ground conductor 1, the second ground conductor 6, the third
ground conductor 7, the first dielectric substrate 8, and the
second dielectric substrate 9 are multilayered; however, a fourth
ground conductor 41 and a third dielectric substrate 42 may be
multilayered as illustrated in FIG. 13.
FIG. 13 is a cross-sectional side view illustrating another antenna
device according to the first embodiment of the invention.
In FIG. 13, the fourth ground conductor 41 is arranged in parallel
with the third ground conductor 7 on, of the two planes of the
third ground conductor 7, a plane opposite to the plane on which
the second ground conductor 6 is arranged.
The third dielectric substrate 42 is arranged between the third
ground conductor 7 and the fourth ground conductor 41.
In the antenna device illustrated in FIG. 13, the portion
sandwiched between the second ground conductor 6 and the third
ground conductor 7 operates as the microstrip resonator 22, and in
addition to this, the portion sandwiched by the third ground
conductor 7 and the fourth ground conductor 41 operates as a
microstrip resonator 43.
Therefore, by adding the fourth ground conductor 41 having sides
the length of which is about half a wavelength at a desired
frequency, a radiation pattern characteristic having a low
cross-polarization can be obtained in multiple frequency bands.
Second Embodiment
In the first embodiment, the example in which the planar shape of
the second ground conductor 6 is a square is illustrated.
In a second embodiment, an example in which notches are formed in
each of the four sides of a second ground conductor 6 as
illustrated in FIG. 14 will be described.
FIG. 14 is a plan view illustrating the planar shape of the second
ground conductor 6 in the antenna device according to the second
embodiment of the present invention. In FIG. 14, the same symbol as
that in FIGS. 1 to 3 represents the same or a corresponding
part.
In the example of FIG. 14, a coaxial line 10 is arranged at the
center of the second ground conductor 6.
Symbols X1, X2, X3, and X4 represent the dimensions of the
respective sides of the second ground conductor 6, and X1=X2=X3=X4
holds.
Symbols Y1, Y2, Y3, and Y4 indicate the notch size of the sides of
the second ground conductor 6.
Where Y1<X1, Y2<X4, Y3<X4, Y4<X1, and Y1=Y2=Y3=Y4
hold.
The second ground conductor 6 having a square planar shape is
notched with the same notch size at the center of each of the four
sides.
Specifically, in FIG. 14, the dimensions of the notches of the
upper side on the paper (hereinafter referred to as the upper
side), the lower side on the paper (hereinafter referred to as the
lower side), the left side on the paper (hereinafter referred to as
the left side), and the right side on the paper (hereinafter
referred to as the right side), of the second ground conductor 6,
are all X2+X3.
Furthermore, the notch sizes on the upper side, the lower side, the
left side, and the right side of the second ground conductor 6 are
all Y (=Y1=Y2=Y3=Y4).
Therefore, even with the notches, the planar shape of the second
ground conductor 6 maintains symmetry, and thus the axial ratio
characteristic can be maintained.
Since the path of a signal flowing through the second ground
conductor 6 becomes longer when a notch is formed in each of the
four sides of the second ground conductor 6, the operation
frequency of the microstrip resonator 22 shifts to the lower
frequency side.
The resonance frequency can be adjusted by adjusting the notch size
Y on the upper side, the lower side, the left side, and the right
side of the second ground conductor 6. Therefore, the phase
relationship can be adjusted not only by the arrangement and the
shapes of the element antennas 3a, 3b, 3c, and 3d but also by
modifying the shape of the second ground conductor 6 by the notches
when the phase relationship between the circularly polarized wave
transmitting/receiving unit 2 that is a current source and the
microstrip resonator 22 that is a magnetic current source is
adjusted.
In the second embodiment, the example in which Y1=Y2=Y3=Y4 holds
for the notch sizes is illustrated in order to maintain the axial
ratio characteristic and to prevent cross polarized waves from
increasing.
In a case where there is no particular problem even if some cross
polarized waves increase due to asymmetry, the notch sizes may be,
for example, Y1.noteq.Y2.noteq.Y3.noteq.Y4. Moreover,
(X2+X3).noteq.(X1+X4) may be satisfied.
Although the example is illustrated in the second embodiment in
which each of the four sides of the second ground conductor 6 is
notched; however, each of the four sides of the third ground
conductor 7 may be notched.
Third Embodiment
In the first embodiment, the example is illustrated in which the
element antennas 3a, 3b, 3c, and 3d are arranged on the first plane
1a of the first ground conductor 1.
In a third embodiment, an example will be described in which third
dielectric substrates 51 arranged on a first plane 1a of a first
ground conductor 1 are further included, and element antennas 3a,
3b, 3c, and 3d are formed in the third dielectric substrates
51.
FIG. 15 is a cross-sectional side view illustrating an antenna
device according to a third embodiment of the invention.
FIG. 16 is a plan view illustrating a top surface of the antenna
device according to the third embodiment of the present
invention.
In FIGS. 15 and 16, the same symbol as that in FIGS. 1 to 3
represents the same or a corresponding part and thus description
thereof is omitted.
The third dielectric substrates 51 are dielectric substrates
stacked on the first plane 1a of the first ground conductor 1 so as
to surround a coaxial line 10.
Inside the third dielectric substrates 51, the element antennas 3a,
3b, 3c, and 3d are formed.
Also, in the case where the element antennas 3a, 3b, 3c, and 3d are
formed inside the third dielectric substrates 51, an antenna device
that operates in a similar manner to the first embodiment is
obtained.
Fourth Embodiment
FIG. 13 of the first embodiment illustrates the antenna device
including the fourth ground conductor 41.
In a fourth embodiment, an antenna device will be described in
which communication component circuits 62, including a filter used
for suppressing unwanted waves, an amplifier for amplifying a
signal or the like when satellite communication is performed for
example, are mounted on a fourth ground conductor 41 as illustrated
in FIG. 17.
FIG. 17 is a cross-sectional side view illustrating an antenna
device according to a fourth embodiment of the invention.
In FIG. 17, the same symbol as that in FIGS. 1 to 3 and 13
represents the same or a corresponding part and thus description
thereof is omitted.
Conductive members 61 are provided so as to pass through a third
dielectric substrate 42 and allow the third ground conductor 7 and
the fourth ground conductor 41 to be conductive therebetween.
The multiple conductive members 61 are arranged at positions
surrounding a coaxial line 10 and communication component circuits
62.
The communication component circuits 62 are attached to, out of the
two planes of the fourth ground conductor 41, the plane opposite to
the plane on which the third ground conductor 7 is arranged, and
includes communication components such as filters or amplifiers
used for satellite communication, for example.
A first metal housing 63 is connected to the fourth ground
conductor 41 so as to shield the communication component circuit 62
from the surroundings thereof.
In the antenna device illustrated in FIG. 17, conductive members 61
allow the third ground conductor 7 and the fourth ground conductor
41 to be conductive therebetween, and the first metal housing 63
protects the communication component circuit 62.
Therefore, even when the antenna device is mounted with the
communication component circuits 62, the antenna device itself can
operate in a similar manner to the first embodiment.
Fifth Embodiment
In the fourth embodiment, the antenna device including the first
metal housing 63 is illustrated.
In a fifth embodiment, an antenna device further including a second
metal housing 64 as illustrated in FIG. 18 will be described.
FIG. 18 is a cross-sectional side view illustrating the antenna
device according to the fifth embodiment of the invention.
In FIG. 18, the same symbol as that in FIGS. 1 to 3 and 17
represents the same or a corresponding part and thus description
thereof is omitted.
The second metal housing 64 is arranged so as to surround a first
metal housing 63.
A resin member 65 is filled between the first metal housing 63 and
the second metal housing 64.
In the antenna device illustrated in FIG. 18, the second metal
housing 64 is arranged so as to surround a first metal housing 63,
and the space between the first metal housing 63 and the second
metal housing 64 is filled with a resin member 65, and thus a
microstrip resonator 66 is formed by the first metal housing 63 and
the second metal housing 64.
In this case, if the electrical length between the first metal
housing 63 and the second metal housing 64 is about half a
wavelength at the resonance frequency, the microstrip resonator 66
operates in a similar manner to the microstrip resonator 22.
According to the fifth embodiment, cross polarized waves can be
suppressed also by the microstrip resonator 66 formed by the first
metal housing 63 and the second metal housing 64.
Note that the present invention may include a flexible combination
of the respective embodiments, a modification of any component of
the embodiments, or an omission of any component in the embodiments
within the scope of the present invention.
INDUSTRIAL APPLICABILITY
The present invention is suitable for an antenna device including a
plurality of element antennas.
REFERENCE SIGNS LIST
1: first ground conductor, 1a: first plane, 1b: second plane, 2:
circularly polarized wave transmitting/receiving unit, 3a, 3b, 3c,
3d: element antenna, 4a, 4b, 4c, 4d: feeding point, 5a, 5b, 5c, 5d:
tip, 6: second ground conductor, 7: third ground conductor, 8:
first dielectric substrate, 9: second dielectric substrate, 10:
coaxial line, 11: outer conductor, 12: penetrating member, 13:
conductor, 14: inner conductor, 14a: one end of inner conductor,
14b: the other end of inner conductor, 15: conductive member, 16:
penetrating member, 17: conductor, 18: interface circuit, 19:
180-degree hybrid, 20, 21: 90-degree hybrid, 22: microstrip
resonator, 30: passive element, 41: fourth ground conductor, 42:
third dielectric substrate, 43: microstrip resonator, 51: third
dielectric substrate, 61: conductive member, 62: communication
component circuit, 63: first metal housing, 64: second metal
housing, 65: resin member, 66: microstrip resonator
* * * * *