U.S. patent number 10,965,032 [Application Number 15/864,288] was granted by the patent office on 2021-03-30 for dielectric resonator antenna.
This patent grant is currently assigned to City University of Hong Kong. The grantee listed for this patent is City University of Hong Kong. Invention is credited to Lei Guo, Kwok Wa Leung, Yong Mei Pan.
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United States Patent |
10,965,032 |
Leung , et al. |
March 30, 2021 |
Dielectric resonator antenna
Abstract
A dielectric resonator antenna includes a dielectric resonator
element, a ground plane, and a conductive feeding arrangement. The
ground plane is connected with the dielectric resonator element,
and is operable to generate a first electromagnetic radiation. The
conductive feeding arrangement is operable to generate a second
electromagnetic radiation. During operation, simultaneous
generation of the first electromagnetic radiation and the second
electromagnetic radiation provides a unilateral electromagnetic
radiation.
Inventors: |
Leung; Kwok Wa (Kowloon Tong,
HK), Pan; Yong Mei (New Territories, HK),
Guo; Lei (Kowloon Tong, HK) |
Applicant: |
Name |
City |
State |
Country |
Type |
City University of Hong Kong |
Kowloon |
N/A |
HK |
|
|
Assignee: |
City University of Hong Kong
(Kowloon, HK)
|
Family
ID: |
1000005456460 |
Appl.
No.: |
15/864,288 |
Filed: |
January 8, 2018 |
Prior Publication Data
|
|
|
|
Document
Identifier |
Publication Date |
|
US 20190214732 A1 |
Jul 11, 2019 |
|
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01Q
1/38 (20130101); H01Q 9/0485 (20130101); H01Q
1/2291 (20130101); H01Q 9/0492 (20130101); H01Q
1/48 (20130101) |
Current International
Class: |
H01Q
9/04 (20060101); H01Q 1/22 (20060101); H01Q
1/38 (20060101); H01Q 1/48 (20060101) |
Field of
Search: |
;343/700MS,793,795,846 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
Primary Examiner: Munoz; Daniel
Attorney, Agent or Firm: Renner Kenner Greive Bobak Taylor
& Weber
Claims
The invention claimed is:
1. A dielectric resonator antenna, comprising: a dielectric
resonator element with a body having a wall; a ground plane in the
form of a patch, attached to the wall of the dielectric resonator
element, the ground plane being sized smaller than the wall of the
dielectric resonator and being operable to be fed into and excite
the dielectric resonator element to a dielectric resonator mode to
generate a first electromagnetic radiation; and a conductive
feeding arrangement comprising a feeding probe received in a space
defined by the body and operable to be fed into and excite the
dielectric resonator element to generate a second electromagnetic
radiation different from and complementary to the first
electromagnetic radiation; wherein, during operation, simultaneous
generation of the first electromagnetic radiation and the second
electromagnetic radiation provides a unilateral electromagnetic
radiation.
2. The dielectric resonator antenna of claim 1, wherein the first
electromagnetic radiation is directed to a first direction and the
second electromagnetic radiation is directed to a second direction
substantially perpendicular to the first direction.
3. The dielectric resonator antenna of claim 1, wherein the first
electromagnetic radiation comprises a magnetic dipole.
4. The dielectric resonator antenna of claim 1, wherein the
dielectric resonator mode is TE.sub.111 mode.
5. The dielectric resonator antenna of claim 1, wherein the ground
plane is provided on a dielectric substrate.
6. The dielectric resonator antenna of claim 1, wherein an angular
position or orientation of the ground plane relative to the
dielectric resonator element is adjustable.
7. The dielectric resonator antenna of claim 1, wherein a footprint
of the ground plane is less than 50% of a footprint of the
dielectric resonator element.
8. The dielectric resonator antenna of claim 1, wherein a footprint
of the ground plane is less than 20% of a footprint of the
dielectric resonator element.
9. The dielectric resonator antenna of claim 1, wherein the second
electromagnetic radiation comprises electric dipole.
10. The dielectric resonator antenna of claim 1, wherein the
conductive feeding arrangement is arranged centrally of the
dielectric resonator element.
11. The dielectric resonator antenna of claim 1, wherein the
feeding probe comprises any of: a cylindrical probe, a conical
probe, an inverted conical probe, and a stepped cylindrical
probe.
12. The dielectric resonator antenna of claim 1, wherein the
feeding probe is an inner conductor of a cable.
13. The dielectric resonator antenna of claim 12, wherein the cable
further comprises an outer conductor operably connected with the
ground plane, and the inner and outer conductors are co-axial.
14. The dielectric resonator antenna of claim 1, wherein the
dielectric resonator element comprises a cuboidal body defining a
space therein for at least partly receiving the conductive feeding
arrangement.
15. The dielectric resonator antenna of claim 1, wherein the
conductive feeding arrangement is substantially perpendicular to
the wall of the dielectric resonator element.
16. The dielectric resonator antenna of claim 1, wherein the
conductive feeding arrangement is substantially perpendicular to
the ground plane.
17. The dielectric resonator antenna of claim 1, wherein the
dielectric resonator antenna is arranged to operate at LTE
band.
18. A dielectric resonator antenna array comprising one or more of
the dielectric resonator antenna of claim 1.
19. A wireless communication system comprising one or more of the
dielectric resonator antenna of claim 1.
20. The dielectric resonator antenna of claim 1, wherein the first
electromagnetic radiation comprises a y-directed magnetic dipole
and the second electromagnetic radiation comprises a z-directed
electric dipole.
Description
TECHNICAL FIELD
The invention relates to a dielectric resonator antenna and
particularly, although not exclusively, to a unilaterally radiating
dielectric resonator antenna with a compact configuration.
BACKGROUND
Laterally radiating antenna can direct radiation in the desired
lateral direction and suppress radiation in the opposite direction.
With relatively low backward radiation, laterally radiating antenna
can desirably reduce power waste and diminish interference with
other devices. Therefore, laterally radiating antennas are
desirable for applications where the communication object or
required coverage range is beside the antenna, such as cordless
phones and Wi-Fi routers that are placed in front of a wall.
Problematically, however, existing laterally radiating antenna
structures for unilateral radiation have complex designs, and so
are rather bulky and difficult to make. There is a need to provide
an improved laterally radiating antenna that is particularly
adapted for use in modern wireless communication systems.
SUMMARY OF THE INVENTION
In accordance with a first aspect of the invention, there is
provided a dielectric resonator antenna, comprising: a dielectric
resonator element; a ground plane connected with the dielectric
resonator element, operable to generate a first electromagnetic
radiation; and a conductive feeding arrangement, operable to
generate a second electromagnetic radiation; wherein, during
operation, simultaneous generation of the first electromagnetic
radiation and the second electromagnetic radiation provides a
unilateral electromagnetic radiation. The ground plane refers to an
electrically conductive surface that is connected to ground, and it
does not have to be strictly planar. The first and second
electromagnetic radiations are preferably complementary.
Preferably, the first electromagnetic radiation is directed to a
first direction and the second electromagnetic radiation is
directed to a second direction substantially perpendicular to the
first direction. For example, the first direction may be in the
y-direction (Cartesian coordinates) and the second direction may be
in the z-direction (Cartesian coordinates).
Preferably, the first electromagnetic radiation comprises a
magnetic dipole. The magnetic dipole may be, for example, a
y-directed magnetic dipole (Cartesian coordinates).
Preferably, the ground plane is arranged to excite a dielectric
resonator mode for generation of the first electromagnetic
radiation. The dielectric resonator mode may be TE.sub.111
mode.
Preferably, the ground plane is in the form of a patch. The patch
may be generally flat.
Preferably, the ground plane is provided on a dielectric
substrate.
Preferably, an angular position or orientation of the ground plane
relative to the dielectric resonator element is adjustable, for
steering the unilateral electromagnetic radiation.
Preferably, a footprint of the ground plane is less than 50% of a
footprint of the dielectric resonator element. More preferably, a
footprint of the ground plane is less than 20% of a footprint of
the dielectric resonator element.
Preferably, the second electromagnetic radiation comprises electric
dipole. The electric dipole may be formed by, for example,
z-directed electric monopole mode in the conductive feeding
arrangement.
Preferably, the conductive feeding arrangement is received in the
dielectric resonator element, and optionally, also arranged
centrally of the dielectric resonator element.
Preferably, the conductive feeding arrangement comprises a feeding
probe, which may be in the form any of: a cylindrical probe, a
conical probe, an inverted conical probe, and a stepped cylindrical
probe.
Preferably, the feeding probe is an inner conductor of a cable. The
cable may further comprise an outer conductor operably connected
with the ground plane, and the inner and outer conductors are
co-axial.
Preferably, the dielectric resonator element comprises a cuboidal
body defining a space therein for at least partly receiving the
conductive feeding arrangement. The cuboidal body may include
squared- or rectangular-cross section. The space preferably
corresponds to the shape and form of the conductive feeding
arrangement.
Preferably, the conductive feeding arrangement is substantially
perpendicular to a wall of the dielectric resonator element.
Preferably, the conductive feeding arrangement is or is also
substantially perpendicular to the ground plane. The ground plane
and the wall may be generally parallel.
Preferably, the dielectric resonator antenna is arranged to operate
at LTE band, in particular, the 3.5 GHz LTE band.
In accordance with a second aspect of the invention, there is
provided a dielectric resonator antenna array comprising one or
more of the dielectric resonator antenna of the first aspect.
In accordance with a third aspect of the invention, there is
provided a wireless communication system comprising one or more of
the dielectric resonator antenna of the first aspect.
BRIEF DESCRIPTION OF THE DRAWINGS
Embodiments of the invention will now be described, by way of
example, with reference to the accompanying drawings in which:
FIG. 1 is a schematic diagram illustrating the basic principle of
complementary unilateral antenna;
FIG. 2 is a schematic diagram of a dielectric resonator antenna in
one embodiment of the invention;
FIG. 3A is a schematic diagram of a first antenna arrangement
(Antenna I) of the dielectric resonator antenna of FIG. 2;
FIG. 3B is a schematic diagram of a second antenna arrangement
(Antenna II) of the dielectric resonator antenna of FIG. 2;
FIG. 4A is a plot showing variation of simulated reflection
coefficient (dB) in the first antenna arrangement of FIG. 3A with
frequency (GHz) for different probe length l.sub.p (8.3 mm, 10.3
mm, and 12.3 mm);
FIG. 4B is a plot showing variation of simulated reflection
coefficient (dB) in the first antenna arrangement of FIG. 3A with
frequency (GHz) for different dielectric resonator element height d
(16.5 mm, 19.5 mm, and 22.5 mm);
FIG. 5 is a plot showing variation of simulated reflection
coefficient (dB) in the second antenna arrangement of FIG. 3B with
frequency (GHz);
FIG. 6A is a plot showing simulated resonant E field in the second
antenna arrangement of FIG. 3B at 2.9 GHz;
FIG. 6B is a plot showing simulated resonant H field in the second
antenna arrangement of FIG. 3B at 2.9 GHz;
FIG. 7A is a plot showing simulated radiation pattern in the E
plane (x-z plane) for the first antenna arrangement of FIG. 3A at
3.9 GHz;
FIG. 7B is a plot showing simulated radiation pattern in the H
plane (x-y plane) for the first antenna arrangement of FIG. 3A at
3.9 GHz;
FIG. 7C is a plot showing simulated radiation pattern in the E
plane (x-z plane) for the second antenna arrangement of FIG. 3B at
2.9 GHz;
FIG. 7D is a plot showing simulated radiation pattern in the H
plane (x-y plane) for the second antenna arrangement of FIG. 3B at
2.9 GHz;
FIG. 8 is a photo showing a dielectric resonator antenna in one
embodiment of the invention, fabricated based on the design
illustrated in FIG. 2;
FIG. 9 is a plot showing simulated and measured reflection
coefficients (dB) of the dielectric resonator antenna of FIG. 8 for
different frequencies (GHz);
FIG. 10A is a plot showing simulated and measured radiation pattern
in the E plane (x-z plane) for the dielectric resonator antenna of
FIG. 8;
FIG. 10B is a plot showing simulated and measured radiation pattern
in the H plane (x-y plane) for the dielectric resonator antenna of
FIG. 8;
FIG. 10C is a plot showing simulated 3D radiation pattern (front
view) for the dielectric resonator antenna of FIG. 8;
FIG. 10D is a plot showing simulated 3D radiation pattern (top
view) for the dielectric resonator antenna of FIG. 8;
FIG. 11 is a plot showing simulated and measured antenna gains
(dBi) of the dielectric resonator antenna of FIG. 8 for different
frequencies (GHz);
FIG. 12 is a plot showing simulated and measured front-to-back
ratio (dB) of the dielectric resonator antenna of FIG. 8 for
different frequencies (GHz);
FIG. 13A is a plot showing variation of simulated reflection
coefficient (dB) in the dielectric resonator antenna of FIG. 8 with
frequency (GHz) for different dielectric resonator element height d
(16.5 mm, 19.5 mm, and 22.5 mm);
FIG. 13B is a plot showing variation of simulated antenna gain
(dBi) in the dielectric resonator antenna of FIG. 8 with frequency
(GHz) for different dielectric resonator element height d (16.5 mm,
19.5 mm, and 22.5 mm);
FIG. 13C is a plot showing variation of simulated front-to-back
ratio (dB) in the dielectric resonator antenna of FIG. 8 with
frequency (GHz) for different dielectric resonator element height d
(16.5 mm, 19.5 mm, and 22.5 mm);
FIG. 14 is a schematic diagram of a dielectric resonator antenna in
another embodiment of the invention, wherein the ground patch is
angularly displaced (by displacement .alpha.) when compared with
FIG. 2;
FIG. 15A is a plot showing variation of simulated reflection
coefficient (dB) in the dielectric resonator antenna of FIG. 14
with frequency (GHz) for different angular displacement .alpha.
(0.degree., 45.degree., and 90.degree.);
FIG. 15B is a plot showing simulated radiation pattern in the E
plane (x-z plane) for the dielectric resonator antenna of FIG. 14
at 3.55 GHz for different angular displacement .alpha. (0.degree.,
45.degree., and 90.degree.);
FIG. 15C is a plot showing simulated radiation pattern in the H
plane (x-y plane) for the dielectric resonator antenna of FIG. 14
at 3.55 GHz for different angular displacement .alpha. (0.degree.,
45.degree., and 90.degree.);
FIG. 16A is a plot showing variation of simulated maximum antenna
gain (dBi) and its corresponding frequency (GHz) for the dielectric
resonator antenna of FIG. 14 with the angular displacement .alpha.;
and
FIG. 16B is a plot showing variation of simulated maximum
front-to-back ratio (dB) and its corresponding frequency (GHz) for
the dielectric resonator antenna of FIG. 14 with the angular
displacement .alpha..
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 shows the basic principle of complementary unilateral
antenna. As shown in FIG. 1, the E- and H-plane radiation patterns
of an electric dipole are of ".infin." and "O" shapes respectively;
and the E- and H-plane radiation patterns of an magnetic dipole are
of "O" and ".infin." shapes respectively. In other words, the
electric dipole and magnetic dipole are of complementary radiation
patterns. In this example, a z-directed electric dipole and a
y-directed magnetic dipole have a constructive interference in x
direction and a destructive interference in -x direction (i.e.,
they substantially cancel each other). The net result is a lateral
unidirectional radiation pattern with good front-to-back ratios
(FTBRs) obtained in both radiation planes.
Thematically, the total far field of a pair of orthogonal electric
and magnetic dipoles can be obtained by superimposing their
individual far field because their fields are orthogonal to each
other. In one example, the total E.sub..theta. and E.sub.O
components of a z-directed electric dipole (length l.sub.e, current
amplitude I.sub.e) and a y-directed magnetic dipole (length
l.sub.m, current amplitude I.sub.m) are given by
.times..times..theta..times..times..pi..times..times..times..times..times-
..omega..function..function..times..times..times..times..times..times..tim-
es..times..times..theta..times..times..delta..times..times..times..times..-
times..times..times..times..PHI..times..times..PHI..times..times..pi..time-
s..times..times..times..times..omega..function..times..times..times..delta-
..times..times..times..times..times..times..times..times..theta..times..ti-
mes..times..times..PHI. ##EQU00001## where k=.omega. {square root
over (.mu..sub.0.epsilon..sub.0)} is the wave number and .delta. is
the phase difference of the two currents. When
.eta.l.sub.eI.sub.e=l.sub.mI.sub.m=lI and .delta.=180.degree., the
total fields can be simplified as:
.times..times..theta..times..times..times..times..pi..times..times..times-
..times..times..omega..function..times..times..times..times..times..times.-
.times..theta..times..times..PHI..times..times..PHI..times..times..times..-
times..pi..times..times..times..times..times..omega..function..times..time-
s..times..times..times..times..times..theta..times..times..times..times..P-
HI. ##EQU00002##
According to equations (3) and (4), the co- and cross-polarized
fields of the E-plane (xz-plane, O=0.degree., 180.degree.) and
H-plane (xy-plane, .theta.=90.degree.) are given by:
Co-Polarized Fields:
|E.sub.T.theta.|(E-plane).varies.|H.sub.TO|(H-plane).varies.(sin
.theta.+cos O) (5)
Cross-Polarized Fields:
|E.sub.TO|(E-plane).varies.|H.sub.T.theta.|(H-plane).varies. cos
.theta. sin O (6)
It can be determined from equation (5) that the co-polarized fields
of both planes are maximum in the +x direction but vanish in the -x
direction. As a result, a cardioid-shaped unilateral pattern with a
large front-to-back (F/B) ratio can be obtained. It can be
determined from equation (6) that the cross-polarized fields vanish
in both planes.
The above analysis is based on magnetic and electric dipoles with
ideal behavior. However, in practice, the vanishing fields can be
of finite values (although still relatively small).
FIG. 2 shows a dielectric resonator antenna 200 in one embodiment
of the invention. The antenna 200 generally includes a dielectric
resonator element 202, a ground plane 204 (electrically conductive
surface connected to ground), and a conductive feeding arrangement
206. The ground plane 204 is arranged to generate a first
electromagnetic radiation, preferably in the form of a magnetic
dipole. The conductive feeding arrangement 206 is arranged to
generate a second electromagnetic radiation, preferably in the form
of an electric dipole. The first electromagnetic radiation may be
directed substantially perpendicularly to the second
electromagnetic radiation. During operation, simultaneous
generation of the first electromagnetic radiation and the second
electromagnetic radiation provides a unilateral electromagnetic
radiation, making the antenna 200 a unilateral dielectric resonator
antenna.
The dielectric resonator element 202 has a generally cuboidal body.
The body defines a space for at least partly receiving the
conductive feeding arrangement 206. The space is arranged centrally
of the dielectric resonator element 202.
The ground plane 204 is in the form of a patch, and it is attached
to a base wall 202B of the dielectric resonator element 202,
extending generally parallel to the base wall 202B. In some
embodiment, the ground plane 204 may be provided on a dielectric
substrate (not shown). In the present embodiment, the ground plane
204 is arranged to excite a dielectric resonator mode for
generation of the first electromagnetic radiation. The dielectric
resonator mode may be TE.sub.111 mode. By adjusting the angular
position or orientation of the ground plane 204 relative to the
dielectric resonator element 202, the radiation pattern can be
steered or adjusted. A footprint of the ground plane 204 is
preferably less than 50%, and more preferably less than 20%, of a
footprint of the dielectric resonator element 202.
The conductive feeding arrangement 206 is a feeding probe of
generally cylindrical form. The probe is received in the space
defined by the body of the dielectric resonator element 202. The
probe is arranged substantially perpendicular to both the base wall
202B of the dielectric resonator element 202 and the ground plane
204. The feeding probe 206 is an inner conductor of a cable, which
may further include an outer conductor operably connected with the
ground plane 204. Preferably, the inner and outer conductors of the
cable are coaxial.
In the present embodiment, the electric and magnetic dipoles are
integrated in a single dielectric resonator antenna 200.
As shown in FIG. 2, the dielectric resonator element 202 has a
square cross section with a side length a, height d, and dielectric
constant .epsilon..sub.r. The dielectric resonator element 202 is
excited in the TE.sub.111 mode by a small rectangular conducting
patch (which forms the ground plane 204) with dimensions of length
l and width w. In this example, the TE.sub.111 mode provides the
required equivalent y-directed magnetic dipole.
A feeding probe 206 of length (i.e., height) l.sub.p and radius
r.sub.p is inserted into the dielectric resonator element 202 at
the center to provide the required z-directed electric monopole
mode. An outer conductor coaxial with the probe and belonging to
the same cable as the probe is connected to the ground patch 204.
In the present example, the field of the TE.sub.111 mode changes
with the angular position or orientation (or displacement) of the
ground patch 204, the unilateral radiation pattern can be easily
steered in the horizontal plane by altering the position or
orientation of the patch 204.
To illustrate the operation of the antenna 200, FIGS. 3A and 3B
provides two antenna arrangements of the dielectric resonator
antenna of FIG. 2. FIG. 3A shows the first antenna arrangement
200A, Antenna I, with the ground patch 204 removed. FIG. 3B shows a
second antenna arrangement 200B, Antenna II, with the probe removed
(probe length l.sub.p=0 mm).
FIGS. 4A and 4B show simulated reflection coefficient of Antenna I
for different probe lengths l.sub.p (FIG. 4A) and dielectric
resonator heights d (FIG. 4B). The following parameters are used in
the simulation: .epsilon..sub.r=10, a=29 mm, and r.sub.p=0.45 mm.
The probe length l.sub.p=8.3 mm, 10.3 mm, and 12.3 mm, with d=19.5
mm (FIG. 4A). The dielectric resonator element height d=16.5 mm,
19.5 mm, and 22.5 mm, with l.sub.p=8.3 mm (FIG. 4B). As shown in
FIGS. 4A and 4B, the resonant frequency decreases significantly
from .about.3.9 to 3.1 GHz as l.sub.p increases from 8.3 to 12.3
mm. However, it changes only slightly when d varies. This indicates
that the resonance at 3.9 GHz is associated with the dielectric
resonator-loaded probe (electric dipole mode).
FIG. 5 shows the simulated reflection coefficient of Antenna II. As
shown in FIG. 5, two resonant modes with poor impedance match are
found in Antenna II. The first resonant mode is found at .about.2.9
GHz. FIGS. 6A and 6B show the simulated resonant E-field and
H-field inside the dielectric resonator element. As shown in FIG.
6A, the E-field basically forms a loop but with slight distortion
at the base caused by the patch. As shown in FIG. 6B, the H-field
is mainly directed along the y direction. These results show that
the first resonant mode found at .about.2.9 GHz is the dominant
TE.sub.111.sup.y mode. On the other hand, the second resonant mode
in FIG. 5 was found to be the higher-order TE.sub.211.sup.y mode.
This mode does not contribute to the required equivalent magnetic
dipole mode.
FIGS. 7A to 7D show the simulated radiation patterns of Antennas I
and II, respectively. As shown in FIGS. 7A to 7D, the radiation
patterns of Antennas I and II are similar to those of a z-directed
electric dipole and y-directed magnetic dipole, respectively. Thus,
a unilateral radiation pattern can be obtained by combining
them.
To demonstrate the above embodiment of the invention, a unilateral
dielectric resonator antenna 800 covering 3.5-GHz LTE band was
designed, fabricated, and tested. FIG. 8 shows a photograph of the
prototype of a dielectric resonator antenna 800. This unilateral
dielectric resonator antenna 800 was designed by ANSYS HFSS and
fabricated by using an ECCOSTOCK HiK dielectric material. The
dielectric resonator antenna 800 has parameters of
.epsilon..sub.r=10, a=29 mm, d=19.5 mm, l=11.5 mm, w=7 mm,
r.sub.p=0.45 mm and l.sub.p=8.3 mm, with loss tangent less than
0.002.
In the antenna 800 of FIG. 8, the ground plane 804 (patch) was
fabricated using a piece of conducting adhesive tape. A semi-rigid
coaxial cable 808 is connected to the ground plane 804 (patch),
with its inner conductor (probe) inserted into the center of the
dielectric resonator element 802 and the outer conductor connected
to the patch 804 (ground). A balun is added to the coaxial cable
808 to suppress stray radiation from the cable. In other
embodiments, the ground plane 804 (patch) can be printed on a
dielectric substrate to enhance the mechanical robustness of the
antenna. In this case, it would be necessary to re-optimize the
antenna design for desired unilateral patterns.
Experiments were performed to obtain various parameters and
measurements of the dielectric resonator antenna 800. In the
experiments, the reflection coefficient was measured using an
HP8510C network analyzer, whereas the radiation pattern, antenna
gain, and antenna efficiency were measured with a Satimo Starlab
System.
FIG. 9 shows the simulated and measured reflection coefficients of
the dielectric resonator antenna prototype. As shown in FIG. 9, the
measured 10-dB impedance bandwidth (|S11|<-10 dB) is 28.5%
(2.86-3.81 GHz), which closely follows the simulated result of
27.0% (2.82-3.70 GHz). The small discrepancy is potentially caused
by experimental imperfections and tolerances. The TE.sub.111.sup.y
mode of the dielectric resonator as found from Antenna II remains
at around 2.9 GHz, despite the inclusion of the probe. This is
reasonable in this example because the probe is located at the
central part of the dielectric resonator element 802 where the
E-field of the TE.sub.111.sup.y mode is weak. In other words, the
coupling between the probe and TE.sub.111.sup.y mode is too small
to obtain the probe effect. In this example, however, the probe
frequency is 3.5 GHz, lower than 3.9 GHz as found in Antenna I, due
to the loading of the patch.
It was found that the dielectric resonator antenna is a good
unilateral antenna at 3.55 GHz. At this frequency, both the
TE.sub.111.sup.y and probe modes are not optimal--the former is not
operated at its resonance frequency (2.9 GHz) whereas the latter is
seriously loaded by the patch. Nevertheless, a unilateral radiation
mode can be obtained as long as the conditions of
.eta.l.sub.eI.sub.e=l.sub.mI.sub.m=lI and .delta.=180.degree. as
discussed above are met. The unilateral radiation mode so obtained
would not be ideal (e.g., a finite F/B ratio) because the
TE.sub.111.sup.y mode (magnetic dipole) and probe mode (electric
dipole) are not pure at this frequency.
FIGS. 10A and 10B show the measured and simulated radiation
patterns at 3.55 GHz. As shown in FIGS. 10A and 10B, both the E-
and H-plane patterns are unilateral. The maximum radiation is found
in the +x direction (.theta.=90, O=0.degree.) with a high F/B ratio
of .about.25 dB. The co-polarized fields of both planes are
stronger than their cross-polarized counterparts by more than 30 dB
in the main (+x) direction. Radiation patterns at other frequencies
were also studied. Very stable results were observed across the
entire LTE passband (not shown). FIGS. 10C and 10D show the 3-D
radiation patterns of the antenna. As shown, the power in the +x
direction is much stronger than that in the -x direction, as
expected.
FIG. 11 shows the measured and simulated antenna gains of the
unilateral dielectric resonator antenna. As shown in FIG. 11,
reasonable agreement between the measured and simulated results is
observed. The measured gain is lower than the simulated result
likely due to experimental imperfections. From FIG. 11, it can be
seen that the measured gain varies between 4.43 dBi and 4.94 dBi
over the LTE band.
FIG. 12 shows measured and simulated front-to-back (F/B) ratios of
the dielectric resonator antenna. As shown in FIG. 12, the measured
and simulated F/B ratios have their maximum values of .about.25 dB,
with the measured 15-dB F/B-ratio bandwidth given by 10.9%
(3.39-3.78 GHz). Both measured and simulated F/B ratios are higher
than 15 dB across the LTE band, which again verifies that the
dielectric resonator antenna is a unilateral antenna with optimal
performance. The efficiency of the dielectric resonator antenna was
also measured, and it was found that the efficiency varies between
82% and 93% across the LTE band.
A comprehensive comparison between the unilateral dielectric
resonator antenna in the present embodiment and the previous design
in L. Guo, K. W. Leung, and Y. M. Pan, "Compact unidirectional ring
dielectric resonator antennas with lateral radiation," IEEE Trans.
Antennas Propag., vol. 63, no. 12, pp. 5334-5342, December 2015 is
given in Table I. As shown in the Table, the current dielectric
resonator antenna has a simpler feeding scheme and a more compact
structure, with its bandwidth comparable to those of the previous
design. Instead of using higher-order dielectric resonator modes
(HEM.sub.11.delta.+1, HEM.sub.11.delta.+2) as found in the previous
design, the fundamental TE111 mode is used for the dielectric
resonator antenna of the present embodiment. This increases the
antenna gain by .about.1 dB in the desired lateral direction
because the fundamental mode has a smaller radiation power density
around the boresight direction (.theta.=0.degree.).
TABLE-US-00001 TABLE I Comparison between current unilateral
dielectric resonator antenna and previous design Aver- Feeding
Permittivity & Usable age Antenna Scheme Dimensions Bandwidth*
Gain Original design using both .epsilon..sub.r = 15 ~4% ~3.7 in
Guo et al. the feeding 1.47 .times. 1.20 .times. 0.89 dBi slot and
probe Wideband using both .epsilon..sub.r = 15 ~14% ~3.4 design in
Guo the feeding 2.17 .times. 0.89 .times. 1.63 dBi et al. slot and
probe The present using only .epsilon..sub.r = 10 11% ~4.6
embodiment the feeding 1.08 .times. 1.08 .times. 0.73 dBi probe
*Usable Bandwidth defined as the overlapping bandwidth between the
10-dB impedance passband and 15-dB F/B ratio passband
A parametric study was carried out to characterize the unilateral
dielectric resonator antenna. The effect of dielectric resonator
size was studied. FIG. 13A shows the simulated reflection
coefficient for d=16.5 mm, 19.5 mm, and 22.5 mm. As shown in FIG.
13A, increasing the dielectric resonator size would decrease the
resonance frequencies. FIGS. 13B and 13C shows the corresponding
simulated antenna gain and F/B ratio, respectively. As shown in
FIGS. 13B and 13C, the frequencies of peak gain and F/B ratio shift
downwards as d increases. This trend is consistent with that of the
reflection coefficient. By comparing FIG. 13A with FIG. 13B, it can
be found that the antenna gain increases with improving impedance
match. The F/B ratio (FIG. 13C), however, conversely decreases with
improving match. This is not surprising because the F/B ratio is
mainly dependent on the relative amplitudes and phases of the
magnetic and electric dipoles, not on the impedance match. The
effect of the dielectric resonator sidelength a was also studied
and similar results were observed (not shown)
The effect of the probe length l.sub.p was investigated. It was
found that the frequency of the peak gain and F/B ratio decreases
with an increase of l.sub.p, showing that the operating frequency
of the antenna can be tuned by changing l.sub.p. It was also found
that good F/B ratio and impedance match can be simultaneously
obtained over the frequency range of 3.25-3.89 GHz, with the
antenna bandwidth varying between .about.2.7% and 9.6% as l.sub.p
decreases from 10 to 6 mm.
The effects of the patch length l and width w were also studied. It
was found that they can be used to adjust the impedance match and
F/B ratio of the antenna, with the effect of 1 being much stronger
than that of w.
In one embodiment of the invention, the beam of the antenna can be
steered in the azimuthal plane by changing the angular orientation
or position (or displacement) of the ground patch. FIG. 14 shows a
dielectric resonator antenna with a ground patch 1404 having an
angular displacement .alpha. (compared with that in FIG. 2). The
construction of the dielectric resonator antenna 1400 is the same
as the dielectric resonator antenna 200 of FIG. 2, except for the
angular position of the ground patch 1404. Three cases of
.alpha.=0.degree., 45.degree., and 90.degree. were studied.
FIGS. 15A to 15C show the simulated reflection coefficient and
radiation pattern, respectively. As shown in FIG. 15A, the results
of .alpha.=0.degree., 90.degree. are the same due to symmetry of
the structure. It can also be observed that the reflection
coefficient of .alpha.=450 is very similar to those of
.alpha.=0.degree., 90.degree.. This is desirable because the
steering can be readily made without substantially affecting
matching. With reference to FIGS. 15B and 15C, the horizontal
radiation pattern rotates as a increases but the vertical radiation
pattern remains substantially unchanged. It should be noted that
the maximum radiation direction is always opposite to the ground
patch, i.e., the maximum radiation will occur at O=.alpha. when the
angular displacement is a. Also, the cardioid shape is
substantially maintained during steering.
FIG. 16A shows the simulated maximum gain and its corresponding
frequency as a function of .alpha.. As shown in FIG. 16A, both the
gain and frequency are symmetry about .alpha.=45.degree. due to the
symmetry of the structure. As a increases from 0.degree. to
45.degree., the maximum gain and corresponding frequency only
slightly increase from 5.12 to 5.33 dBi and from 3.47 to 3.52 GHz,
respectively. FIG. 16B shows the simulated maximum F/B ratio and
its corresponding frequency as a function of a. Again, the
variations are very small as a varies. All these results show that
stable cardioid-shaped radiation pattern can be maintained when
doing the steering.
The above embodiments of the invention have provided a simple
laterally radiating rectangular dielectric resonator antenna that
has a feeding probe and a small ground patch. In the illustrated
embodiment, the dielectric resonator element is excited in its
fundamental TE.sub.111 mode to provide an equivalent magnetic
dipole. This magnetic dipole is combined with the electric monopole
of the feeding probe to give a lateral cardioid-shaped radiation
pattern. The unilateral dielectric resonator antennas in the above
embodiments have small ground plane and thus are compact. The
antenna can be simply fed by the inner conductor of a SMA
connector, omitting the need of complex feeding network. The
antenna is largely made of dielectric and so the loss can be made
small even at mm-wave frequencies. This in turn provides high
radiation efficiency. Different bandwidths for different
applications can be obtained, by selecting suitable dielectric
constant to be used in the unilateral dielectric resonator antenna
of the present invention. The lateral radiation pattern of the
dielectric resonator antenna of the above embodiments can be easily
steered in different horizontal directions by changing the angular
position, orientation, or displacement of the ground patch, with no
significant effects on impedance match.
It will be appreciated by persons skilled in the art that numerous
variations and/or modifications may be made to the invention as
shown in the specific embodiments without departing from the spirit
or scope of the invention as broadly described. For example, the
dielectric resonator element can be of any shape, not necessarily
cuboidal. The ground plane can be of any shape and form. The probe
can be of any shape and form, such as a conical probe, an inverted
conical probe, and a stepped cylindrical probe. Any other
dielectric resonator mode can be used to provide the equivalent
magnetic dipole, not necessarily the fundamental TE.sub.111 mode.
The permittivity .epsilon..sub.r of the dielectric resonator
element can be of any value. The present embodiments are,
therefore, to be considered in all respects as illustrative and not
restrictive.
* * * * *