U.S. patent number 10,700,651 [Application Number 16/191,409] was granted by the patent office on 2020-06-30 for wide bandpass filtering power amplifier.
This patent grant is currently assigned to South China University of Technology. The grantee listed for this patent is South China University of Technology. Invention is credited to Qinchuang Chen, Yuanchun Li, Quan Xue.
View All Diagrams
United States Patent |
10,700,651 |
Li , et al. |
June 30, 2020 |
Wide bandpass filtering power amplifier
Abstract
A wide bandpass filtering power amplifier using discriminating
coupling is disclosed, which comprises a DC bias circuit, an input
impedance matching circuit, a transistor and an output impedance
matching circuit. The DC bias circuit is connected to the input
impedance matching circuit which is further connected to the
transistor, and the transistor is further connected to the output
impedance matching circuit which comprises a tuning microstrip line
and a bandpass filter. The complexity and the area of the impedance
matching circuit in the power amplifier are effectively reduced. At
the same time, the filtering PA has good frequency selectivity by
using the discriminating coupling BPF. Meanwhile the work
efficiency and bandwidth of the filtering power amplifier are
effectively improved by taking both of the extended continuous mode
theory and filter synthesis theory into account.
Inventors: |
Li; Yuanchun (Guangdong,
CN), Chen; Qinchuang (Guangdong, CN), Xue;
Quan (Guangdong, CN) |
Applicant: |
Name |
City |
State |
Country |
Type |
South China University of Technology |
Guangzhou, Guangdong |
N/A |
CN |
|
|
Assignee: |
South China University of
Technology (Guangzhou, CN)
|
Family
ID: |
64898191 |
Appl.
No.: |
16/191,409 |
Filed: |
November 14, 2018 |
Prior Publication Data
|
|
|
|
Document
Identifier |
Publication Date |
|
US 20200028478 A1 |
Jan 23, 2020 |
|
Foreign Application Priority Data
|
|
|
|
|
Jul 23, 2018 [CN] |
|
|
2018 1 0815617 |
|
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01P
7/082 (20130101); H03F 3/2171 (20130101); H01P
1/20363 (20130101); H03F 1/0205 (20130101); H03F
2200/171 (20130101); H03F 2200/387 (20130101); H03F
2200/222 (20130101); H03F 2200/423 (20130101) |
Current International
Class: |
H03F
3/217 (20060101); H03F 1/02 (20060101); H01P
7/08 (20060101) |
Field of
Search: |
;330/302,305-306,251,277 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Nguyen; Hieu P
Claims
What is claimed is:
1. A wide bandpass filtering power amplifier comprising a DC bias
circuit, an input impedance matching circuit, a transistor and an
output impedance matching circuit, wherein the DC bias circuit is
connected to the input impedance matching circuit which is further
connected to the transistor, and the transistor is further
connected to the output impedance matching circuit which comprising
a tuning microstrip line and a bandpass filter, wherein the tuning
microstrip line is connected between the transistor and the
bandpass filter; wherein the bandpass filter comprises a first
resonator and a second resonator parallelly coupled with each
other; wherein the first resonator comprises a first short end and
a first open end, and the second resonator comprises a second short
end and a second open end, wherein the first resonator has a
coupling region of 2/3 L1 starting from the first short end and the
second resonator has a coupling region of 2/3 L2 starting from the
second open end, wherein L1 represents a length of the first
resonator and L2 represents a length of the second resonator.
2. The wide bandpass filtering power amplifier according to claim
1, wherein L1=L2=.lamda..sub.g/4, wherein .lamda..sub.g represents
a waveguide wavelength at working frequency.
3. The wide bandpass filtering power amplifier according to claim
1, the first short end of the first resonator is aligned with the
second open end of the second resonator.
4. The wide bandpass filtering power amplifier according to claim
1, the first short end of the first resonator is connected to a DC
power source, and is further grounded via a capacitor.
5. The wide bandpass filtering power amplifier according to claim
1, wherein an input terminal of the bandpass filter is connected to
the first resonator, and an output terminal of the bandpass filter
is connected to the second resonator.
6. The wide bandpass filtering power amplifier according to claim
1, wherein an input terminal of the tuning microstrip line is
connected to the drain of the transistor, an output terminal of the
tuning microstrip line is connected to an input terminal of the
bandpass filter whose output terminal is matched to a load
impedance.
7. The wide bandpass filtering power amplifier according to claim
1, wherein a length of the tuning microstrip line depends on an
imaginary part of an optimal fundamental impedance matching
point.
8. The wide bandpass filtering power amplifier according to claim
1, wherein the input impedance matching network of the bandpass
filter is a Chebyshev bandpass filter.
9. The wide bandpass filtering power amplifier according to claim
1, wherein the transistor is a GaN HEMT CGH40010F transistor.
10. A wide bandpass filtering power amplifier comprising a DC bias
circuit, an input impedance matching circuit, a transistor and an
output impedance matching circuit, wherein the DC bias circuit is
connected to the input impedance matching circuit which is further
connected to the transistor, and the transistor is further
connected to the output impedance matching circuit which comprising
a tuning microstrip line and a bandpass filter, wherein the tuning
microstrip line is connected between the transistor and the
bandpass filter; wherein the bandpass filter comprises a first
resonator and a second resonator parallelly coupled with each
other; wherein the first resonator comprises a first microstrip
line, a second microstrip line, a third microstrip line and a
fourth microstrip line connected sequentially, and the second
resonator comprise a fifth microstrip line, a sixth microstrip
line, a seventh microstrip line and an eighth microstrip line
connected sequentially.
11. The wide bandpass filtering power amplifier according to claim
10, wherein the second microstrip line, the third microstrip line
and the fourth microstrip line of the first resonator are
respectively parallelly coupled with the fifth microstrip line, the
sixth microstrip line and the seventh microstrip line of the second
resonator.
12. The wide bandpass filtering power amplifier according to claim
11, wherein the second microstrip line, the third microstrip line
and the fourth microstrip line of the first resonator are connected
in a direct line, while the fifth micro strip line, the sixth
microstrip line and the seventh microstrip line of the second
resonator are also connected in a direct line.
13. The wide bandpass filtering power amplifier according to claim
12, wherein the first microstrip line and the second microstrip
line of the first resonator are vertically connected with each
other, while the seventh microstrip line and the eighth microstrip
line of the second resonator are connected with each other in a
direct line.
14. The wide bandpass filtering power amplifier according to claim
10, wherein an input terminal of the bandpass filter is connected
between the third microstrip line and the fourth microstrip line,
and an output terminal of the bandpass filter is connected between
the seventh microstrip line and the sixth microstrip line.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
The present application claims the benefits of Chinese Patent
Application No. 2018108156177, filed on Jul. 23, 2018, the content
of which is incorporated herein by reference in its entirety.
TECHNICAL FIELD
The present disclosure relates generally to technical fields of a
power amplifier, and more particularly, to a wide bandpass
filtering power amplifier.
BACKGROUND
With the development of communication technology, the standard of
wireless communication system has the characteristics of high data
rate, large user capacity and low power consumption. As a key
component of front-end, power amplifier has a great influence on
transmitter performance. Therefore, the power amplifier with high
efficiency and broadband operation has made considerable
progress.
To improve efficiency, a variety of switch mode PAs are proposed,
such as Class-D, Class-E and Class-F PAs. Theoretically, the
efficiency of the above power amplifier can reach 100%. Doherty and
envelope tracking structure are used to improve efficiency and
linearity. To maintain bandwidth efficiency and maintain high
efficiency, various technologies have been put forward. On the
basis of standard switch mode PAs, continuous Class-B/J, continuous
Class-F and continuous Class-F.sup.-1, are proposed, which
alleviate the requirement on terminating impedance of harmonics
from fixed values into purely reactive regions and thus achieve
satisfactory bandwidth. After that, extended continuous mode PA
further relaxes the requirements of harmonic impedances to
reactive-resistive ones with a small degradation in efficiency.
Power amplifiers are usually cascaded with low insertion loss
bandpass filters to suppress the out-of-band interference. Although
the bandwidth of the continuous power amplifier has reached one
octave, the overall bandwidth and efficiency are degraded due to
the interconnection mismatching and insertion loss of the filter.
In order to overcome this problem, the co-design of power amplifier
and bandpass filter has been studied widely. They mainly focus on
miniaturization, power efficiency enhancement (PAE) and selectivity
improvement.
However, the preceding filtering PA designs have high selectivity
and overall PAE, while the bandwidth with large PAE is relatively
small due to the limited design freedom.
SUMMARY
The object of the present application is to provide a wide bandpass
filtering power amplifier which takes both of the extended
continuous mode theory and filter synthesis theory into account.
The work efficiency and bandwidth of the filtering power amplifier
are effectively improved, while the complexity and the area of the
power amplifier are effectively reduced.
In one aspect, a wide bandpass filtering power amplifier is
provided, comprising a DC (direct current) bias circuit, an input
impedance matching circuit, a transistor and an output impedance
matching circuit. The DC bias circuit is connected to the input
impedance matching circuit which is further connected to the
transistor. Meanwhile the transistor is connected to the output
impedance matching circuit which comprises a tuning microstrip line
and a bandpass filter, wherein the tuning microstrip line is
connected between the transistor and the bandpass filter (BPF).
In a preferable embodiment of the present application, the bandpass
filter comprises a first resonator and a second resonator
parallelly coupled with each other.
In the wide bandpass filtering power amplifier according to the
present application, the first resonator comprises a first short
end and a first open end, and the second resonator comprises a
second short end and a second open end, wherein the first resonator
has a coupling region of 2/3 L1 starting from the first short end
and the second resonator has a coupling region of 2/3 L2 starting
from the second open end, wherein L1 represents a length of the
first resonator and L2 represents a length of the second
resonator.
In a preferable embodiment of the present application, the first
short end of the first resonator is aligned with the second open
end of the second resonator. In a further preferable embodiment of
the present application, L1=L2=.lamda..sub.g/4, wherein
.lamda..sub.g represents a waveguide wavelength at working
frequency. In a further preferable embodiment of the present
application, the first short end of the first resonator is
connected to a DC power source, and is further grounded via a
capacitor.
In the wide bandpass filtering power amplifier according to the
present application, an input terminal of the bandpass filter is
connected to the first resonator, and an output terminal of the
bandpass filter is connected to the second resonator.
In the wide bandpass filtering power amplifier according to the
present application, an input terminal of the tuning microstrip
line is connected to a drain of the transistor, an output terminal
of the tuning microstrip line is connected to an input terminal of
the bandpass filter whose output terminal is matched to a load
impedance.
In the wide bandpass filtering power amplifier according to the
present application, a length of the tuning microstrip line depends
on an imaginary part of an optimal fundamental impedance matching
point.
In the wide bandpass filtering power amplifier according to the
present application, the first resonator comprises a first
microstrip line, a second microstrip line, a third microstrip line
and a fourth microstrip line connected sequentially, and the second
resonator comprise a fifth microstrip line, a sixth microstrip
line, a seventh microstrip line and an eighth microstrip line
connected sequentially.
In a preferable embodiment of the present application, the second
microstrip line, the third microstrip line and the fourth
microstrip line of the first resonator are respectively parallelly
coupled with the fifth microstrip line, the sixth microstrip line
and the seventh microstrip line of the second resonator. In a
further preferable embodiment of the present application, the
second microstrip line, the third microstrip line and the fourth
microstrip line of the first resonator are connected in a direct
line, while the fifth microstrip line, the sixth microstrip line
and the seventh microstrip line of the second resonator are also
connected in a direct line. In a further preferable embodiment of
the present application, the first microstrip line and the second
microstrip line of the first resonator are vertically connected
with each other, while the seventh microstrip line and the eighth
microstrip line of the second resonator are connected with each
other in a direct line.
In a further preferable embodiment of the present application, an
input terminal of the band-pass filter is connected between the
third microstrip line and the fourth microstrip line, and an output
terminal of the band-pass filter is connected between the seventh
microstrip line and the sixth microstrip line.
In the wide bandpass filtering power amplifier according to the
present application, the input impedance matching network of the
bandpass filter is a Chebyshev bandpass filter.
In the wide bandpass filtering power amplifier according to the
present application, the transistor is a GaN HEMT CGH40010F
transistor.
By the implementation of the wide bandpass filtering power
amplifier, several advantages can be obtained. The complexity and
the area of the impedance matching circuit in the conventional
Class-F.sup.-1 power amplifier are effectively reduced by
integrating the DC bias circuit into the bandpass filter, while
satisfying the selectivity of the PA. Meanwhile the work efficiency
and bandwidth of the filtering power amplifier are effectively
improved by taking the extended continuous mode theory and filter
synthesis theory into account to guide the impedance matching
design of the wide bandpass filtering power amplifier.
BRIEF DESCRIPTION OF THE DRAWINGS
To illustrate more clearly the technical scheme of the embodiments
of the present application, the accompanying drawings to be used in
the description of the present embodiment will be briefly described
below. The drawings described below are only some embodiments of
the present application, and for one skilled in the art other
drawings may be obtained from them without creative effort.
FIG. 1 is a block diagram of a wide bandpass filtering power
amplifier using discriminating coupling according to an embodiment
of the present application.
FIG. 2 is a circuit diagram of a discriminating coupling bandpass
filtering output matching network of power amplifier according to
an embodiment of the present application.
FIG. 3 is a schematic diagram of the theoretical normalized voltage
and current waveforms at the drain of an ideally extended
continuous Class-F.sup.-1 power amplifier.
FIG. 4 is a schematic diagram of the voltage distributions in a
bandpass filter with discriminating coupling according to an
embodiment of the present application.
FIG. 5 shows measured and simulated S-parameters of the wide
bandpass filtering power amplifier using discriminating coupling
according to an embodiment of the present application.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
These and other advantage, aspect and novel features of the present
application, as well as details of an illustrated embodiment
thereof, will be more fully understand from the following
description and drawings. Apparently, the described embodiments are
some embodiments of the present application, rather than all of
them. Based on the embodiments of the present application, all
other embodiments acquired by one skilled in the art without
creative work shall fall within the scope of the present
application.
It should be noted that, the terms "first", "second", "third" and
"fourth" in the descriptions, claims and the appended drawings of
the present application are used to distinguish different objects,
rather than to describe a particular order. In addition, the terms
"include", "have", "comprise", the likes, and any variants of them,
are intended to cover any other possible inclusions.
Hereinafter, embodiments of the present application will be
described in detail with reference to the accompanying
drawings.
FIG. 1 is a block diagram of a wide bandpass filtering power
amplifier using discriminating coupling according to an embodiment
of the present application. As shown in FIG. 1, the wide bandpass
filtering power amplifier using discriminating coupling comprises a
DC (direct current) bias circuit 20, an input impedance matching
circuit 10, a transistor 30 and an output impedance matching
circuit 40. The DC bias circuit 20 is connected to the input
impedance matching circuit 10 which is further connected to the
transistor 30. Meanwhile the transistor 30 is connected to the
output impedance matching circuit 40 which comprise a tuning
microstrip line 410 and a bandpass filter 420, wherein the tuning
microstrip line 410 is connected between the transistor 30 and the
bandpass filter 420. The bandpass filter 420 comprises a first
resonator 422 and a second resonator 424 parallelly coupled with
each other.
To be specific, the input terminal circuit of the transistor 30 is
consisted of the DC bias circuit 20 and the input impedance
matching circuit 10, for improving the roll-off at the passband
edge. In the present embodiment, the mentioned DC bias circuit 20
and the input impedance matching circuit 10 can be the prior DC
bias circuit and input impedance matching circuit used in the prior
conventional Class-F.sup.-1 power amplifier, and are not described
in the present application for concise.
To be specific, in the present application, the output terminal
circuit of the transistor 30 consists of the tuning microstrip line
410 and the bandpass filter 420 with harmonic suppression and
harmonic control functions. In the present application, the
bandpass filter 420 has harmonic suppression, harmonic control as
well as DC biasing functions. The bandpass filter 420 consists of
the first resonator 422 and the second resonator 424 parallelly
coupled with each other and both having a length of
.lamda..sub.g/4, wherein .lamda..sub.g represents a waveguide
wavelength at working frequency. Accordingly, the second harmonic
and the fourth harmonic can be suppressed.
To be specific, in the present application, the bandpass filter 420
can be a Chebyshev bandpass filter. In a more preferable
embodiment, the transistor 30 can be a GaN HEMT CGH40010F
transistor from Cree.
Furthermore, as shown in FIG. 1, an input terminal of the tuning
microstrip line 410 is connected to a drain of the transistor 30,
an output terminal of the tuning microstrip line 410 is connected
to an input terminal of the bandpass filter 420 whose output
terminal is matched to a load impedance. The tuning microstrip line
has an impedance conversion function. The optimal fundamental
impedance matching point located at the transistor package plane
can be converted to be an impedance with a small imaginary part,
which is then matched with the load impedance via the bandpass
filter.
FIG. 2 is a circuit diagram of a wide bandpass filtering power
amplifier using discriminating coupling according to an embodiment
of the present application. As shown in FIG. 2, the first resonator
422 comprises a first short end and a first open end, and the
second resonator 424 comprises a second short end and a second open
end. The first resonator 422 has a coupling region of 2/3 L1
starting from the first short end and the second resonator 424 has
a coupling region of 2/3 L2 starting from the second open end,
wherein L1 represents the length of the first resonator 422 and L2
represents the length of the second resonator 424. The first short
end of the first resonator 422 is aligned with the second open end
of the second resonator 424. In a further preferable embodiment of
the present application, L1=L2=.lamda..sub.g/4, wherein
.lamda..sub.g represents the waveguide wavelength at working
frequency. In this way, the third harmonic can be suppressed by
reasonably choosing the coupling regions of the two resonators, so
that the bandpass filter can suppress the second, third and fourth
harmonics and provide the second harmonic and the third harmonic
control function.
Furthermore, as shown in FIG. 2, the first short end of the first
resonator 422 is connected to a DC power source 426, and is further
grounded via a capacitor 428. Thus, while providing harmonic
suppression and harmonic control, the bandpass filter also has a DC
biasing function by connecting the short end with the DC power
source.
Furthermore, as shown in FIG. 2, the input terminal of the bandpass
filter 420 is connected to the first resonator 422, and the output
terminal of the bandpass filter 420 is connected to the second
resonator 424. The detail connection position can be adjusted
according to the input matching and the output matching of the
bandpass filter 420. In this way, the impedance condition of the
extended continuous Class-F.sup.-1 power amplifier can be satisfied
by adjusting the length of the resonator of the bandpass filter,
the connection position of the input and output terminals and the
distance between the two resonators.
Furthermore, as shown in FIG. 2, the first resonator 422 (referring
FIG. 1) comprises a first microstrip line 1, a second microstrip
line 2, a third microstrip line 3 and a fourth microstrip line 4
connected sequentially, and the second resonator 424 (referring
FIG. 1) comprise a fifth microstrip line 5, a sixth microstrip line
6, a seventh microstrip line 7 and an eighth microstrip line 8
connected sequentially. In the present embodiment, the second
microstrip line 2, the third microstrip line 3 and the fourth
microstrip line 4 of the first resonator 422 are respectively
parallelly coupled with the fifth microstrip line 5, the sixth
microstrip line 6 and the seventh microstrip line 7 of the second
resonator 424.
The input terminal of the bandpass filter 420 is connected between
the third microstrip line 3 and the fourth microstrip line 4, and
the output terminal of the bandpass filter 420 is connected between
the seventh microstrip line 7 and the sixth microstrip line 6. It
should be noted that, the input terminal of the bandpass filter 420
can be arranged at different locations on the first, second, third
and fourth microstrip lines 1-4, and meanwhile the output terminal
of the bandpass filter 420 can also be arranged at different
locations on the fifth, sixth, seventh and eight microstrip lines
5-8, and both of which can be adjusted according to the output
matching impedance.
In a further preferable embodiment of the present application, the
second microstrip line 2, the third microstrip line 3 and the
fourth microstrip line 4 of the first resonator 422 are connected
in a direct line, while the fifth microstrip line 5, the sixth
microstrip line 6 and the seventh microstrip line 7 of the second
resonator 424 are also connected in a direct line. In such a way,
the second microstrip line 2, the third microstrip line 3 and the
fourth microstrip line 4 are parallelly coupled with the fifth
microstrip line 5, the sixth microstrip line 6 and the seventh
microstrip line 7. It should be noted that, there can be different
connection manners between the first microstrip line 1 and the
second microstrip line 2, and between the seventh microstrip line 7
and the eighth microstrip line 8, so long as the first microstrip
line 1 and the eighth microstrip line 8 are not parallelly coupled.
FIG. 2 has shown a propositional structure, in which the first
microstrip line 1 and the second microstrip line 2 of the first
resonator 422 are vertically connected with each other, while the
seventh microstrip line 7 and the eighth microstrip line 8 of the
second resonator 424 are connected with each other in a direct
line. Through such kinds of structures, the coupling region of the
two resonators can be guaranteed to have the third harmonic
suppression effect.
The working principle of the wide bandpass filtering power
amplifier using discriminating coupling provided by the present
application is analyzed in detail below.
FIG. 3 is a schematic diagram of the theoretical normalized voltage
and current waveforms at the drain of an ideally extended
continuous Class-F.sup.-1 (CCF.sup.-1) power amplifier, which is
the wide bandpass filtering power amplifier using discriminating
coupling provided by the present application. The standard
Class-F.sup.-1 mode PA needs
the optimum impedance for the fundamental signal, as well as a
constant open-circuit for the second harmonic and short-circuit for
the third harmonic. However, the harmonic impedance of the output
matching network (OMN) changes at the edge of Smith Chart and
should be difficult to be kept as zero or infinite for different
harmonics, which limits the bandwidth of high-efficiency PA. The
extended CCF.sup.-1 mode PA with resistive second harmonic
impedance is put forward for expanding the impedance region in the
Smith Chart.
The normalized drain voltage of the extended CCF.sup.-1 mode PA is
the same as that of the standard Class-F.sup.-1 mode, which is
described as in (1)
.function..theta..times..times..theta..times..times..times..theta..times.-
.times..times..times..times..theta. ##EQU00001##
On the basis of the conventional Class-F.sup.-1 amplifier's current
expression, the parameters .gamma. and .alpha. are introduced to
shape the waveforms. Equation (2) represents the normalized drain
current as follows: i.sub.ds(.theta.)=(i.sub.DC-i.sub.1 cos
.theta.+i.sub.3 cos 3.theta.)(1-.gamma. sin .theta.)(1+.alpha. cos
.theta.)-1.ltoreq..gamma..ltoreq.1. (2) where, i.sub.DC=0.37,
i.sub.1=0.43, i.sub.3=0.06.
As shown in FIG. 3, the normalized voltage and current waveforms of
the extended CCF.sup.-1 mode are depicted, in which it can be found
that the voltage waveform maintains a half-sinusoidal function.
However, when .alpha. increases, the amplitude of the current
waveform decreases, which results in a light deterioration of
output power and drain efficiency. The normalized optimal
admittance could be calculated through equation (3) as follows:
##EQU00002##
However, the operating bandwidth of the transistor is increased
while the efficiency and output power are maintained through the
expanding of the impedance condition of the extended continuous
Class-F.sup.-1 power amplifier of the present application. To
achieve this ideal efficiency, the nth normalized harmonics load
admittances are derived in (4) as follows:
.times..times..alpha..times..times..gamma..function..times..alpha..times.-
.alpha..times..times..gamma..function..times..alpha..infin.
##EQU00003##
In which, G.sub.opt is the optimal conductance of the fundamental
wave and Y.sub.n is the nth harmonic admittance. The output
matching network matches the harmonics to the corresponding
impedance conditions on the current generator surface.
Accordingly, the design flexibility is expanded and the second
harmonic admittance is not limited in a purely reactive region as
before. Moreover, .alpha. is determined to be in the range from 0
to 0.4 to guarantee the efficiency higher than 70% by calculating
the drain efficiency. Each harmonic admittance of the Output
Matching Network should present in the corresponding regions of
Smith Chart.
However, as shown in FIG. 2, there are many parasitic components
between the output pin and the actual drain of the internal chip,
such as parasitic inductance L.sub.out and parasitic capacitance
C.sub.out during the actual applications. Therefore, when designing
the output matching network, the influence of parasitic components
should be taken into account, while the specific values of the
parasitic inductance and parasitic capacitance can be obtained from
the producer.
The BPF with discriminating coupling is used as OMN to make the
fundamental and second harmonic impedances located in the high
efficiency regions, respectively for designing wideband high
efficiency PA with a filtering response, as the OMN is the most
important part in PA design which determines the output power,
efficiency and bandwidth.
FIG. 4 is a schematic diagram of a bandpass filter with
discriminating coupling according to an embodiment of the present
application. Here, G.sub.opt=0.053 is chosen and Y.sub.n in
previous section is calculated to provide the continuous modes
operations with the required admittance.
As shown in FIG. 4, the OMN is just consisted of a tuning
microstrip line and a BPF with two .lamda..sub.g/4 resonators 1 and
2 discriminatively coupled to each other. It can be found that the
OMN is concise in configuration and compact in size. In this
embodiment, the first resonator 1 has a coupling region of 2/3 L
starting from the first short end and the second resonator 2 has a
coupling region of 2/3 L starting from the second open end, wherein
L represents the lengths of the first resonator 1 and the second
resonator 2 which are both equal to .lamda..sub.g/4, wherein
.lamda..sub.g represents a waveguide wavelength at working
frequency. The first short end of the first resonator 1 is aligned
with the second open end of the second resonator 2.
The BPF exits the fundamental passband and the third harmonic
spurious passband as the .lamda..sub.g/4 resonators are employed in
this embodiment. The overall coupling strength at each mode can be
described by the sum of electric coupling coefficient k.sub.e and
magnetic coupling coefficient k.sub.m. On the microstrip
resonators, the dominant mode is quasi-TEM and the electric
coupling coefficient is expressed as follows:
.times..intg..function..times..function..times..intg..function..times..ti-
mes..intg..function..times. ##EQU00004##
wherein V.sub.1 and V.sub.2 are the voltage distributions on the
lines within the coupling regions of the first resonator 1 and the
second resonator 2 and p is a constant. The voltages at the open
end and short end of the resonator are the maximum and zero,
respectively. The normalized voltages at 3f.sub.0 on the first
resonator 1 V.sub.R.sub.1.sub.,3f.sub.0 and on the second resonator
2 V.sub.R.sub.2.sub.,3f.sub.0 are expressed, respectively as
follows: V.sub.R.sub.1.sub.,3f.sub.0=cos [3.beta..sub.0(x-L)] (6);
V.sub.R.sub.2.sub.,3f.sub.0=cos(3.beta..sub.0x). (7);
where .beta..sub.0 is the propagation constant at f.sub.0. By
substituting equations (6) and (7) into equation (5), the coupling
strength at 3f.sub.0 is:
.times..times..times..intg..times..times..times..times..function..times..-
times..function..times..intg..times..times..times..times..function..times.-
.times..intg..times..times..times..times..function..times..times..intg..ti-
mes..times..times..function..times..times..beta..function..times..function-
..times..times..beta..times..times..intg..times..times..times..function..t-
imes..times..beta..function..times..times..intg..times..times..times..func-
tion..times..times..beta..times..times. ##EQU00005##
As in equation (8), V.sub.R.sub.1.sub.,3f.sub.0 is an even function
while V.sub.R.sub.2.sub.,3f.sub.0 is an odd function in the
coupling region. The integrand in the numerator is zero, which
represents the electric coupling coefficient k.sub.e,3f.sub.0
should be zero, likewise k.sub.m,3f.sub.0 also should be zero.
Therefore, the total coupling strength k.sub.(1,2),3f.sub.0 should
zero at the third harmonic.
The coupling matrix of the filter is used for calculating the input
impedance of the third harmonic and the coupling matrix of a
second-order BPF is expressed as follows:
.times..times..times..times..function..times..times..times..times..times.-
.times..times. ##EQU00006##
Accordingly, as shown in FIG. 2, the input impedance at node A can
be expressed as follows:
.function..times..times..times..times..times..times..function..times..tim-
es..times..times. ##EQU00007##
In this equation, as M.sub.12,3f.sub.0=k.sub.(1,2),3f.sub.0/FBW,
M.sub.12,3f.sub.0 equals to zero at 3f.sub.0, and meanwhile, the
lengths of the two resonators are the same guaranteeing
discriminating coupling. Thus, M.sub.11,3f.sub.0 and
M.sub.22,3f.sub.0 have the same values.
Then, the input impedance at 3f.sub.0 is simplified as follows:
.times..times..times..times..times. ##EQU00008##
The values of M.sub.11,3f.sub.0 and M.sub.22,3f.sub.0 can be
adjusted accordingly by tuning the length of the resonators
slightly. It should be noted that, M.sub.S1,3f.sub.0 is related
with the input position, and an arbitrary reactance can be obtained
at node A. The short-circuit at the current generator plane (I-gen
plane) can be realized in a relatively easy way by the mentioned
design. From the above analysis, it can be concluded that the third
harmonic is shorted at the I-gen plane without requiring any extra
circuit by using the discriminating coupling. The requirement of
the extended CCF.sup.-1 PA is fulfilled.
The equation (4) can be used for calculating the impedance region
in the Smith Chart of the second harmonic. The calculated second
impedances at the intrinsic I-gen plane are converted to the
required second harmonic impedances at the package plane for design
convenience. The input impedance at the second harmonic
Z.sub.B,2f.sub.0 at a proper value must be design to fulfill the
requirement.
As the second harmonic is suppressed by the 4/4 resonator filter
intrinsically, the input impedance Z.sub.A,2f.sub.0 at 2f.sub.0 is
a reactive value, and Z.sub.B,2f.sub.0 is calculated as
follows:
.times..times..times..times..times..times..times..theta..times..times..ti-
mes..times..times..times..theta..times. ##EQU00009##
where Z.sub.T is the characteristic impedance of the tuning
microstrip line. Just because the extended CCF.sup.-1 mode is
employed, the required impedance of the second harmonic has a large
design freedom which provides great freedom of Z.sub.T and
.theta..sub.T,2f.sub.0.
It should be noted that the second harmonic impedance condition of
the extended CCF.sup.-1 PA can be extended on the current
generating surface by adjusting the characteristic impedance and
length of the tuning microstrip line. Meanwhile, it also should be
noted that the difficulty of matching the fundamental impedance of
the filter can be reduced by converting the fundamental complex
impedance at point A into an impedance at point B with a smaller
imaginary through adjusting the length of the tuning microstrip
line.
The DC bias circuit 20 is simply connected at the short-end of the
first resonator, and in such a way that the bias scheme can
terminate the second harmonic in the defined region when comparing
with the conventional bias.
The fundamental impedance conversion is analyzed by the coupling
matrix of the BPF at f.sub.p. When designing the wideband filtering
PA, a low Q-factor BPF in the OMN is required. It is found that the
desired BPF response has 30% fractional bandwidth (FBW) and 0.14 dB
insertion loss, and the initial coupling matrix with the impedance
normalized to 1.OMEGA. is synthesized as follows:
##EQU00010##
Starting with M.sub.11=M.sub.22=0, the normalized input impedance
at f.sub.0 can be expressed by equation (18)
.times..times..times..times..times. ##EQU00011##
M.sub.S1, M.sub.11 and M.sub.22 are modified as M.sub.S1'M.sub.11'
and M.sub.22' for performing the complex impedance conversion, and
then the coupling matrix with complex input impedance is modified
as follows:
.times..times.'.times..times.'''.times..times..times.
##EQU00012##
The modified normalized input impedance can be expressed as
follows:
'.times..function..times..times..times..times.''.function.'.times..times.-
.times..times.'.times..times..function..times.' ##EQU00013##
Assuming M.sub.11-M.sub.22, the modified M.sub.S1'M.sub.11' and
M.sub.22' are calculated by solving the equation (20).
The impedance region of fundamental signal at package plane is
converted from I-gen plane at 1.8 GHz with
-1.ltoreq..gamma..ltoreq.1 and 0.ltoreq..alpha..ltoreq.0.4, and the
normalized Z.sub.B,f.sub.0=(0.37+j0.41).OMEGA. is selected in the
package plane region. Moreover, the tuning microstrip line
transforms the desired complex load impedance to an impedance with
a small imaginary part for filtering in an easy way, then
Z.sub.A,f.sub.0'=(0.33+j0.18).OMEGA. is obtained. The modified
coupling matrix can be expressed as follows:
##EQU00014##
When the value of Z.sub.A,f.sub.0' is converted to 50.OMEGA. by the
discriminating coupling filter in the output matching network, the
coupling coefficient and the external quality factors of the BPF
should fulfill the matrix, and the coupling coefficient
K.sub.(1,2),f.sub.0, between the first resonator 1 and the second
resonator 2 as well as the external quality factors
Q.sub.ein,f.sub.0' and Q.sub.eout,f.sub.0 are calculated by
equations (21)-(22):
.times.'.times..times..times.'.times..times..times..times..times..times.
##EQU00015##
Accordingly, the three values of k.sub.(1,2),f.sub.0,
Q.sub.ein,f.sub.0' and Q.sub.eout,f.sub.0 are as follows:
k.sub.(1,2),f.sub.0=0.39, Q.sub.ein,f.sub.0'=2.21, and
Q.sub.eout,f.sub.0=3.10.
According to the coupling coefficient equation (5), the coupling
strength k.sub.(1,2),f.sub.0, of the BPF at f.sub.0 is:
.times..intg..times..times..times..function..times..function..times..intg-
..times..times..times..function..times..times..intg..times..times..times..-
function..times..noteq. ##EQU00016##
Where, V.sub.R.sub.1.sub.,f.sub.0 and V.sub.R.sub.2.sub.,f.sub.0
represent the fundamental voltage distributions of Resonator 1 and
Resonator 2 in FIG. 4, respectively. Although the coupling
coefficient at 3f.sub.0 is zero, the coupling coefficient at
f.sub.0 is not. The desired value of k.sub.(1,2),f.sub.0, can be
modulated by optimizing the gap, and meanwhile, the desired values
of Q.sub.ein,f.sub.0' and Q.sub.eout,f.sub.0 can also be adjusted
by modifying the input positions of feeding lines. Once the
required values of the k.sub.(1,2),f.sub.0, Q.sub.ein,f.sub.0' and
Q.sub.eout,f.sub.0 are satisfied, the required impedance condition
and filtering response are also satisfied.
FIG. 5 shows measured and simulated S-parameters of the wide
bandpass filtering power amplifier using discriminating coupling
according to an embodiment of the present application. A preselect
Chebyshev BPF is utilized as the input matching network (IMN) for
improving the roll-off at the passband edges, and obtaining the
input matching.
By the implementation of the wide bandpass filtering power
amplifier using discriminating coupling, several advantages can be
obtained. The complexity and the area of the filtering power
amplifier are effectively reduced while possessing the frequency
selectivity. Meanwhile the work efficiency and bandwidth of the
filtering power amplifier are effectively improved by taking both
of the extended continuous mode theory and filter synthesis theory
into account to guide the design of the wide bandpass filtering
power amplifier.
To sum up, the present application relates to wide bandpass
filtering power amplifier using discriminating coupling which is an
extended CCF.sup.-1 mode PA integrated with a discriminating
coupling BPF, and the technical solution and effect of the extended
CCF.sup.-1 mode and the OMN using discriminating coupling filter
has been analyzed. The impedances of fundamental mode and the third
harmonic have been converted to the desired values independently as
the discriminating coupling is employed. At the same time, as the
DC voltage is supplied through the BPF, it helps the second
harmonic located in the wanted impedance region in the Smith Chart.
The wide bandpass filtering power amplifier using discriminating
coupling according to the present application has compact size,
good frequency selectivity and high PAE of 73.5%. More importantly,
it is further noted that the wide bandpass filtering power
amplifier using discriminating coupling according to the present
application processes 31.1% FBW with the PAE larger than 60% and it
would be useful in the miniaturized digital transmitters with the
relative wide PAE bandwidth.
The foregoing is a further detailed description of the present
application in connection with specific preferred embodiments, and
cannot be considered as that the specific implementation of the
present application is limited to these illustrations. It will be
apparent to those skilled in the art that any various modifications
or substitutions may be made to the present application without
departing from the spirit of the application, and such
modifications or substitutions should be considered as falling
within the scope of the present application.
* * * * *