U.S. patent number 10,270,380 [Application Number 15/751,183] was granted by the patent office on 2019-04-23 for power converting apparatus and heat pump device.
This patent grant is currently assigned to Mitsubishi Electric Corporation. The grantee listed for this patent is Mitsubishi Electric Corporation. Invention is credited to Kazunori Hatakeyama, Takahiko Kobayashi, Yosuke Shinomoto, Keisuke Uemura, Yasuhiko Wada.
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United States Patent |
10,270,380 |
Uemura , et al. |
April 23, 2019 |
Power converting apparatus and heat pump device
Abstract
A power converting apparatus includes: an inverter converting a
direct-current voltage supplied from a power supply unit into an
alternating-current voltage and outputting the alternating-current
voltage to a motor; and an inverter control unit outputting
synchronous PWM (Pulse Width Modulation) signals for driving
switching elements of the inverter. A frequency of the synchronous
PWM signals is periodically varied when periodic pulsation occurs
in a load connected to the motor.
Inventors: |
Uemura; Keisuke (Tokyo,
JP), Hatakeyama; Kazunori (Tokyo, JP),
Shinomoto; Yosuke (Tokyo, JP), Kobayashi;
Takahiko (Tokyo, JP), Wada; Yasuhiko (Tokyo,
JP) |
Applicant: |
Name |
City |
State |
Country |
Type |
Mitsubishi Electric Corporation |
Tokyo |
N/A |
JP |
|
|
Assignee: |
Mitsubishi Electric Corporation
(Tokyo, JP)
|
Family
ID: |
58186856 |
Appl.
No.: |
15/751,183 |
Filed: |
September 4, 2015 |
PCT
Filed: |
September 04, 2015 |
PCT No.: |
PCT/JP2015/075211 |
371(c)(1),(2),(4) Date: |
February 08, 2018 |
PCT
Pub. No.: |
WO2017/037945 |
PCT
Pub. Date: |
March 09, 2017 |
Prior Publication Data
|
|
|
|
Document
Identifier |
Publication Date |
|
US 20180241336 A1 |
Aug 23, 2018 |
|
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
F25B
31/02 (20130101); H02P 27/085 (20130101); H02M
7/53871 (20130101); H02P 21/05 (20130101); Y02B
30/70 (20130101); F25B 2600/021 (20130101); F25B
13/00 (20130101) |
Current International
Class: |
H02P
27/08 (20060101); F25B 31/02 (20060101); H02M
7/5387 (20070101); H02P 21/05 (20060101); F25B
13/00 (20060101) |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
|
|
|
|
|
|
|
2007-244106 |
|
Sep 2007 |
|
JP |
|
2007244106 |
|
Sep 2007 |
|
JP |
|
2011-066949 |
|
Mar 2011 |
|
JP |
|
2011061884 |
|
Mar 2011 |
|
JP |
|
Other References
International Search Report of the International Searching
Authority dated Oct. 6, 2015 for the corresponding International
application No. PCT/JP2015/075211 (and English translation). cited
by applicant .
Office action dated Aug. 28, 2018 issued in corresponding JP patent
application No. 2017-537180 (and English machine translation
thereof). cited by applicant.
|
Primary Examiner: Dinh; Thai T
Attorney, Agent or Firm: Posz Law Group, PLC
Claims
The invention claimed is:
1. A power converting apparatus comprising: an inverter converting
a direct-current voltage supplied from a power supply unit into an
alternating-current voltage and outputting the alternating-current
voltage to a motor; and an inverter control unit outputting a
synchronous PWM (Pulse Width Modulation) signal for driving a
switching element of the inverter, wherein a frequency of the
synchronous PWM signal periodically varies with a change in a rotor
position of the motor when periodic pulsation occurs in a load
connected to the motor, and the frequency of the synchronous PWM
signal is periodically varied on a basis of an electric current
flowing to the motor when periodic pulsation occurs in the load
connected to the motor.
2. A heat pump device comprising the power converting apparatus
according to claim 1.
Description
CROSS REFERENCE TO RELATED APPLICATION
This application is a U.S. national stage application of
International Patent Application No. PCT/JP2015/075211 filed on
Sep. 4, 2015, the disclosure of which is incorporated herein by
reference.
TECHNICAL FIELD
The present invention relates to a power converting apparatus that
converts a direct-current voltage into an alternating-current
voltage, and also relates to a heat pump device.
BACKGROUND
Synchronous Pulse Width Modulation (PWM) signals for driving
switching elements that constitute an inverter are often generated
by a PWM method. The PWM method is a method to appropriately
control the ratio between ON-time and OFF-time in a switching
period that is a specific period of time in order to output a
voltage that can be instantaneously output by the switching
elements, that is, a voltage between a voltage applied between a
collector and an emitter and a zero voltage, as the average voltage
in the switching period.
The PWM method includes an asynchronous PWM method and a
synchronous PWM method. A synchronous PWM signal for a switching
element in the PWM method is generated by a carrier-wave comparison
method to compare the magnitude of an intended output-voltage
command value and a carrier wave. In the following descriptions,
the intended output-voltage command value is referred to as
"output-voltage command value" and the carrier wave is referred to
as "carrier".
The synchronous PWM method is a method to control the carrier
frequency so as to become an integer multiple of the frequency of
an output-voltage command value, that is, to control the carrier
frequency on the basis of the frequency of the output-voltage
command value. The asynchronous PWM method is a method used when
the carrier frequency is fixed at any frequency of the
output-voltage command value.
The frequency of the output-voltage command value varies on the
basis of the load state. In particular, in a case where a load of
an inverter is a motor, as the rotation speed of the motor is
increased, the frequency of the output-voltage command value
becomes higher. Patent Literature 1 discloses an inverter device to
which direct-current power is input. By using the PWM method, the
inverter device applies alternating-current power of a given
voltage and frequency to a motor to drive the motor at variable
speeds.
Patent Literature
Patent Literature 1: Japanese Patent Application Laid-open No.
2011-66949
Furthermore, Patent Literature 1 discloses controlling the carrier
frequency so as to become an integer multiple of the frequency of
an output-voltage command value and controlling the amount of
change in a PWM frequency on the basis of the motor acceleration
and noise level.
However, Patent Literature 1 describes an asynchronous PWM method
to control the carrier frequency and the voltage command value
asynchronously and does not take any measure against pulsation of
the load. When the inverter device described in Patent Literature 1
is applied to a compressor of an air conditioner, the load torque
may pulsate periodically due to intake, compression, and discharge
in an operating cycle of the compressor, and accordingly the
rotation speed and the rotation position of the motor may pulsate.
Thus, the inverter device cannot sometimes perform synchronous PWM
modulation in a stable manner.
SUMMARY
The present invention has been achieved to solve the above
problems, and an object of the present invention is to provide a
power converting apparatus that can perform synchronous PWM
modulation in a stable manner even when the load torque pulsates
periodically.
To solve the above problems and achieve the object, a power
converting apparatus according to an aspect of the present
invention includes: an inverter converting a direct-current voltage
supplied from a power supply unit into an alternating-current
voltage and outputting the alternating-current voltage to a motor;
and an inverter control unit outputting a synchronous PWM (Pulse
Width Modulation) signal for driving a switching element of the
inverter. A frequency of the synchronous PWM signal is periodically
varied when periodic pulsation occurs in a load connected to the
motor.
A power converting apparatus according to the present invention
produces an effect where synchronous PWM modulation can be
performed in a stable manner even when the load torque pulsates
periodically.
BRIEF DESCRIPTION OF DRAWINGS
FIG. 1 is a diagram illustrating a configuration of a power
converting apparatus according to a first embodiment.
FIG. 2 is a diagram illustrating a configuration of a motor control
unit and a load-pulsation compensating unit according to the first
embodiment.
FIG. 3 is a diagram illustrating a relation between voltage command
values and a carrier frequency according to the first
embodiment.
FIG. 4 is a diagram illustrating a configuration of a synchronous
PWM control unit and a compensating unit according to the first
embodiment.
FIG. 5 is a diagram illustrating a relation between a load torque
and a motor rotation speed according to the first embodiment.
FIG. 6 is a diagram illustrating a relation between a rotor
rotation axis and a control axis according to the first
embodiment.
FIG. 7 is a diagram illustrating a relation between a load torque,
an actual rotor rotation speed, a velocity command value, a
velocity command-value compensation amount, and a carrier-frequency
command-value compensation amount according to the first
embodiment.
FIG. 8 is a diagram illustrating a relation between a load torque,
an actual rotor rotation speed, and an actual rotor rotation
position according to a second embodiment.
FIG. 9 is a diagram illustrating a configuration of a motor control
unit and a load-pulsation compensating unit according to the second
embodiment.
FIG. 10 is a diagram illustrating a configuration of a synchronous
PWM control unit and a compensating unit according to the second
embodiment.
FIG. 11 is a diagram illustrating a relation between a load torque,
an actual rotor rotation speed, and a q-axis current according to a
third embodiment.
FIG. 12 is a diagram illustrating a configuration of a motor
control unit and a load-pulsation compensating unit according to
the third embodiment.
FIG. 13 is a diagram illustrating a configuration of a synchronous
PWM control unit and a compensating unit according to the third
embodiment.
FIG. 14 is a diagram illustrating a configuration of a heat pump
device according to a fourth embodiment.
FIG. 15 is a diagram illustrating a configuration example of
hardware for realizing the power converting apparatus according to
the first to third embodiments.
DETAILED DESCRIPTION
A power converting apparatus and a heat pump device according to
embodiments of the present invention will be described in detail
below with reference to the accompanying drawings. The present
invention is not limited by the embodiments.
First Embodiment
FIG. 1 is a diagram illustrating a power converting apparatus 1
according to a first embodiment of the present invention. The power
converting apparatus 1 includes an inverter 12 that converts a
direct-current voltage supplied from a direct-current power supply
11 that is a power supply unit into an alternating-current voltage
and outputs the alternating-current voltage to a motor 2; an
inverter control unit 13 that outputs synchronous PWM signals for
driving switching elements of the inverter 12; a
direct-current-voltage detecting unit 14 that detects a voltage Vdc
of the direct-current power supply 11; and a current detecting unit
15 that detects an electric current Idc flowing to the inverter
12.
The direct-current power supply 11 may be configured such that an
alternating-current voltage is rectified by using a diode bridge to
be converted into a direct-current voltage and the converted
direct-current voltage is smoothed by using a smoothing capacitor.
Furthermore, the direct-current power supply 11 may be configured
from a direct-current power supply, a representative example of
which is a solar cell or a battery.
The inverter 12 is configured to include switching elements 16a,
16b, 16c, 16d, 16e, and 16f and diodes 17a, 17b, 17c, 17d, 17e, and
17f connected in parallel with the switching elements 16a, 16b,
16c, 16d, 16e, and 16f, respectively.
Each of the switching elements 16a, 16b, 16c, 16d, 16e, and 16f is
constituted by a transistor, an Insulated Gate Bipolar Transistor
(IGBT), a Metal Oxide Semiconductor-Field Effect Transistor
(MOS-FET), a thyristor, or a Gate Turn-Off Thysistor (GTO). The
switching elements 16a, 16b, 16c, 16d, 16e, and 16f may be made of
a semiconductor material such as Si or a wide bandgap semiconductor
material such as SiC or GaN.
On the basis of the voltage Vdc detected by the
direct-current-voltage detecting unit 14 and the electric current
Idc detected by the current detecting unit 15, the inverter control
unit 13 generates synchronous Pulse Width Modulation (PWM) signals
UP, VP, WP, UN, VN, and WN, and applies the generated synchronous
PWM signals UP, VP, WP, UN, VN, and WN to the inverter 12.
Specifically, the synchronous PWM signal UP is applied to the
switching element 16a. The synchronous PWM signal VP is applied to
the switching element 16b. The synchronous PWM signal WP is applied
to the switching element 16c. The synchronous PWM signal UN is
applied to the switching element 16d. The synchronous PWM signal VN
is applied to the switching element 16e. The synchronous PWM signal
WN is applied to the switching element 16f.
In the inverter 12, on the basis of the application of the
synchronous PWM signals UP, VP, WP, UN, VN, and WN, the switching
elements 16a, 16b, 16c, 16d, 16e, and 16f are driven to apply a
given voltage to the motor 2. The motor 2 is driven on the basis of
the applied voltage.
The inverter control unit 13 includes a motor control unit 18 that
generates voltage command values; a synchronous PWM control unit 19
that generates the synchronous PWM signals UP, VP, WP, UN, VN, and
WN; and a load-pulsation compensating unit 20 that generates
signals for compensating for pulsation of a load.
The configuration and operation of the motor control unit 18 are
described below. FIG. 2 is a diagram illustrating the configuration
of the motor control unit 18 and the load-pulsation compensating
unit 20 according to the first embodiment. The motor control unit
18 includes a current restoring unit 21 that restores an electric
current; a converting unit 22 that converts three phase currents
into two phase currents and performs d-q conversion on the two
phase currents; an estimating unit 23 that estimates the position
and the velocity, a velocity control unit 24 that controls the
velocity; a current control unit 25 that controls an electric
current; and a voltage-command calculating unit 26 that generates
voltage command values.
The current restoring unit 21 restores phase currents Iu, Iv, and
Iw flowing to the motor 2 on the basis of the electric current Idc
detected by the current detecting unit 15.
On the basis of a rotor magnetic-pole position .theta. of the motor
2, the converting unit 22 converts the phase currents Iu, Iv, and
Iw, which are three phase currents, into two phase currents, and
converts the two phase currents into a d-axis current Id and a
q-axis current Iq on d-q coordinate axes.
The estimating unit 23 calculates the rotor magnetic-pole position
.theta. and an estimated velocity value .omega. of the motor 2 on
the basis of the electric currents Id and Iq and voltage command
values Vd* and Vq* generated by the current control unit 25.
The velocity control unit 24 calculates a q-axis current command
value Iq* such that the estimated velocity value .omega.
corresponds with a velocity command value .omega.*.
The current control unit 25 calculates a d-axis voltage command
value Vd* such that the d-axis current Id corresponds with an
externally-input d-axis current command value Id*, and calculates a
q-axis voltage command value Vq* such that the q-axis current Iq
corresponds with the q-axis current command value Iq*.
The voltage-command calculating unit 26 calculates UVW-phase
voltage command values Vu*, Vv*, and Vw* on the basis of the d-axis
voltage command value Vd*, the q-axis voltage command value Vq*,
the voltage Vdc detected by the direct-current-voltage detecting
unit 14, and the rotor magnetic-pole position .theta..
FIG. 3(a) is a diagram illustrating a relation between the carrier
frequency and the UVW-phase voltage command values Vu*, Vv*, and
Vw* generated by the voltage-command calculating unit 26. FIG. 3(b)
is a diagram illustrating waveforms of the synchronous PWM signals
UP, VP, WP, UN, VN, and WN generated by the synchronous PWM control
unit 19.
The voltage-command calculating unit 26 generates a voltage phase
.theta.v and outputs the generated voltage phase .theta.v to the
synchronous PWM control unit 19. Specifically, the voltage-command
calculating unit 26 outputs the voltage phase .theta.v using a
falling zero-cross point of Vu* as a reference point. That is, the
reference point is represented as "voltage phase .theta.v=0". Any
point may be used as a reference point of the voltage phase
.theta.v.
The synchronous PWM control unit 19 compares the carrier and the
UVW-phase voltage command values Vu*, Vv*, and Vw* and generates
the synchronous PWM signals UP, VP, WP, UN, VN, and WN, which will
be described later in detail.
Next, the configuration and operation of the load-pulsation
compensating unit 20 are described. The load-pulsation compensating
unit 20 includes a compensating unit 31 that calculates a velocity
command-value compensation amount .DELTA..omega.* and a
carrier-frequency command-value compensation amount .DELTA.fc*; and
an adder 32 that generates the velocity command value .omega.*.
The compensating unit 31 calculates the velocity command-value
compensation amount .DELTA..omega.* and the carrier-frequency
command-value compensation amount .DELTA.fc* on the basis of a
velocity command value .omega.*.sub.(ave) provided from a
higher-level controller for the motor control unit 18. The
compensating unit 31 compensates for the velocity command value
.omega.*.sub.(ave) by using the velocity command-value compensation
amount .DELTA..omega.*. Further, the compensating unit 31
compensates for the carrier by using the carrier-frequency
command-value compensation amount .DELTA.fc*. The velocity
command-value compensation amount .DELTA..omega.* and the
carrier-frequency command-value compensation amount .DELTA.fc* will
be described later in detail.
The adder 32 adds the velocity command value .omega.*.sub.(ave) and
the velocity command-value compensation amount .DELTA..omega.*
together to generate the velocity command value .omega.*.
Next, the configuration and operation of the synchronous PWM
control unit 19 are described. FIG. 4 is a diagram illustrating the
configuration of the synchronous PWM control unit 19 according to
the first embodiment. The synchronous PWM control unit 19 includes
a carrier generating unit 33 that generates a carrier; and a
carrier comparing unit 34 that generates the synchronous PWM
signals UP, VP, WP, UN, VN, and WN.
The carrier generating unit 33 generates a carrier to be
synchronized with the voltage phase .theta.v generated by the
voltage-command calculating unit 26. Further, the carrier
generating unit 33 compensates for the carrier by using the
carrier-frequency command-value compensation amount .DELTA.fc*. The
carrier-frequency command-value compensation amount .DELTA.fc* will
be described later in detail.
The carrier generating unit 33 controls the frequency of a
triangular-wave carrier so as to become 3n times the frequency of
the U-phase voltage command value Vu*, where n is a natural number
equal to or larger than one. The carrier generating unit 33 may
control the frequency of a triangular-wave carrier so as to become
3n times the frequency of the V-phase voltage command value Vv* or
the frequency of the W-phase voltage command value Vw*.
The carrier comparing unit 34 compares the carrier and the voltage
command value Vu* and outputs a "High" or "Low" synchronous PWM
signal. In a case where the frequency of a triangular-wave carrier
is three times the frequency of a voltage command value, a
synchronous PWM signal includes three pulses. In a case where the
frequency of a triangular-wave carrier is six times the frequency
of a voltage command value, a synchronous PWM signal includes six
pulses. In a case where the frequency of a triangular-wave carrier
is nine times the frequency of a voltage command value, a
synchronous PWM signal includes nine pulses.
In a case where the carrier frequency is set to be nine or more
times the frequency of a voltage command value, the number of
pulses of a synchronous PWM signal per period of the voltage
command value is increased. This improves the output-voltage
accuracy. However, because the number of times the switching
elements 16a, 16b, 16c, 16d, 16e, and 16f are switched increase,
the switching loss is increased. That is, there is a trade-off
relation between the magnitude of the carrier frequency and the
switching loss.
Thus, the power converting apparatus 1 compensates for the velocity
command value .omega.*.sub.(ave) by using the velocity
command-value compensation amount .DELTA..omega.* generated by the
load-pulsation compensating unit 20. Also, the power converting
apparatus 1 compensates for the carrier by using the
carrier-frequency command-value compensation amount .DELTA.fc*
generated by the load-pulsation compensating unit 20. Thus, when
periodic pulsation has occurred in a load connected to the motor 2,
the frequency of a synchronous PWM signal output from the
synchronous PWM control unit 19 is varied periodically such that
the power converting apparatus 1 can perform synchronous PWM
modulation in a stable manner.
A description will be given below of the influence on the rotation
speed of the motor and on the rotation phase of the rotor when a
load in which torque pulsation occurs at a constant period is
connected to the motor 2 and of the cases when the response speed
of the velocity control unit 24 is high and when the response speed
of the velocity control unit 24 is low.
FIG. 5 is a diagram illustrating a relation between the load torque
and the actual rotor rotation speed in a case where the load in
which torque pulsation occurs at a constant period is connected to
the motor 2. In the following descriptions, the load is a
compressor in which the load torque pulsates per period in
mechanical angle. However, a pulsation pattern of the load torque
illustrated in FIG. 5 is merely an example and any pulsation
pattern may be employed.
When the velocity command value .omega.* is constant, the velocity
control unit 24 generates the q-axis current command value Iq* that
is a motor-current command value so as to maintain the motor
rotation speed constant. It is known that the motor rotation speed
is represented by the following Equation (1). .omega..sub.m
indicates the actual rotor rotation speed that is the actual motor
angular velocity. .tau..sub.m indicates the motor output torque.
.tau..sub.l indicates the load torque. J.sub.m indicates the moment
of inertia of the motor and the load.
d.omega..sub.m/dt-(.tau..sub.m-.tau..sub.l)/J.sub.m (1)
On the basis of Equation (1), where ".tau..sub.m>.tau..sub.l",
the rotor of the motor 2 is brought into an accelerating state,
while where ".tau..sub.m<.tau..sub.l", the rotor of the motor 2
is brought into a decelerating state.
Thus, in order to control the rotor rotation speed of the motor 2
to be constant, it is necessary to control the motor output torque
such that ".tau..sub.m=.tau..sub.l" is satisfied. In particular,
when the moment of inertia J.sub.m is lower, the value on the right
side of Equation (1) becomes larger. Thus, the sensitivity of the
motor 2 to the rotor speed is increased. In order to control the
motor output torque such that ".tau..sub.m=.tau..sub.l" is
satisfied, it is necessary to increase the response speed of the
velocity control unit 24. There is thus a possibility of causing
overshoot in a transient response at the time of a start-up
operation of the motor 2.
Further, when the response speed of the velocity control unit 24 is
increased, the response speed of the current control unit 25 that
is a minor loop also needs to be increased. It is general that the
response speed of the current control unit 25 needs to be ten or
more times the response speed of the velocity control unit 24. This
may increase a harmonic component of a motor-current waveform and
thus high-frequency sound may be generated from the motor 2.
Furthermore, when the response speed of the velocity control unit
24 and the current control unit 25 is increased, it is necessary to
increase the estimated velocity value .omega., a low-pass filter
time constant .tau..sub..omega., for the d-axis current Id, and a
low-pass filter time constant .tau..sub.i for the q-axis current
Iq.
In an actual motor driving device to which the power converting
apparatus 1 is applied, a pulsating component may be generated in
the estimated velocity value .omega., the d-axis current Id, and
the q-axis current Iq. In particular, an offset that does not occur
theoretically may cause an offset in the electric current Idc
detected by the current detecting unit 15. In that case, in the
current restoring unit 21, an offset occurs also in the phase
currents Iu, Iv, and Iw having been restored on the basis of the
electric current Idc. Further, an offset is superimposed as a
pulsating component on the d-axis current Id and the q-axis current
Iq that have been converted into a rotating coordinate system by
the converting unit 22.
Because the frequency of this pulsating component is proportional
to the rotation speed of the motor 2, a low-pass filter time
constant (a cutoff frequency) is set on the basis of the minimum
rotation speed of the motor 2. In order to ensure the stability of
the velocity control unit 24 and the current control unit 25, the
relation between the low-pass filter time constant (the cutoff
frequency) and the response speed of the velocity control unit 24
and the current control unit 25 is important.
In order to increase the response speed in the entire control
system, it is necessary to increase the low-pass filter time
constant .tau..sub..omega., for the d-axis current Id and the
low-pass filter time constant .tau..sub.i for the q-axis current
Iq. In that case, in a low motor-speed range, the pulsating
component cannot be completely removed from the estimated velocity
value .omega., the d-axis current Id, and the q-axis current Iq.
Thus, the entire control system may become unstable and the
rotation speed of the motor 2 also may become unstable.
It is general that motor control calculation is performed in
synchronization with a carrier. A control period is determined by
the carrier frequency. Thus, in order to increase the response
speed of the entire control system, it is necessary to increase the
control frequency as well as the carrier frequency. However, when
the carrier frequency is increased, the number of times the
inverter 12 is switched is increased and this increases the
switching loss. Further, the level of vibration and radiation noise
generated from the inverter 12 may be increased.
Next, a case where the response speed of the velocity control unit
24 is reduced is described. When the response speed of the velocity
control unit 24 is reduced, the actual rotor rotation speed
.omega..sub.m may become unstable. Moreover, follow-up performance
of a voltage output from the inverter 12 may be degraded, pulsation
may occur in the actual rotor rotation speed .omega..sub.m, and
noise may be generated from the motor 2.
Further, when the response speed of the velocity control unit 24 is
reduced, a phase difference may occur between the rotor
magnetic-pole position .theta. calculated by the estimating unit 23
and an actual rotor rotation position .theta..sub.1 of the motor 2,
and current ripples may occur on the basis of the phase
difference.
FIG. 6 is a diagram illustrating a relation between the actual
rotor rotation position .theta..sub.1 and the rotor magnetic-pole
position .theta. of the motor 2. In the following descriptions, the
d-q coordinates are defined on the actual rotor rotation position
of the motor 2. The coordinates estimated by the estimating unit 23
in the control are defined on .gamma.-.delta. axes. Further, in a
case of sensorless position control, the motor control unit 18 does
not have a mechanism that directly detects the d-q axes and thus
executes control on the .gamma.-.delta. axes estimated by the
estimating unit 23. However, as illustrated in FIG. 6,
.gamma.-.delta. coordinates do not always correspond with the d-q
coordinates.
In particular, in a case where pulsation occurs in a torque load,
the actual rotor rotation speed .omega..sub.m pulsates and
accordingly the actual rotor rotation position .theta..sub.1 also
pulsates. However, in a case where the response speed of the
velocity control unit 24 is low, a phase relation .theta..sub.v
between a voltage vector v output from the inverter 12 and the
actual rotor rotation position .theta..sub.1 pulsates and thus
ripples are generated in the phase currents Iu, Iv, and Iw.
When ripples are generated in the phase currents Iu, Iv, and Iw,
the noise level from the motor 2 may be increased. Further, a motor
driving method may deviate from a motor driving method that is
intended by a control-system designer and achieve maximum
efficiency control or maximum power-factor control.
Furthermore, when the response speed of the velocity control unit
24 is reduced, there is a possibility of interference with the
synchronous PWM control. In the synchronous PWM control, the
carrier frequency and the frequency of a voltage command value are
controlled so as to have a fixed relationship, and thereby the
inverter 12 and the motor 2 are controlled in a stable manner even
at a low carrier frequency.
Thus, if the carrier frequency and the frequency of the voltage
command value cannot be controlled so as to have a fixed
relationship, a frequency component that is different from an
intended frequency component is superimposed on a PWM signal for
driving a switching element of the inverter 12 and on a voltage
command value output from the inverter 12 to the motor 2. In
particular, when the number of pulses in the synchronous PWM
control is set small, there is a tendency for the sensitivity to
the current ripples to increase, the current ripples being caused
by the phase difference between the carrier and the voltage command
value output from the inverter 12 to the motor 2 and also caused by
the phase difference between the actual rotor rotation position
.theta..sub.1 and the rotor magnetic-pole position .theta. of the
motor 2.
The current ripples make the motor control and the synchronous PWM
control unstable. Thus, the phase difference between the carrier
and the voltage command value output from the inverter 12 to the
motor 2 and the phase difference between the actual rotor rotation
position .theta..sub.1 and the rotor magnetic-pole position .theta.
of the motor 2 may increase.
Accordingly, in a case where a load in which torque pulsation
occurs at a constant period is connected to the motor 2, the power
converting apparatus 1 according to the first embodiment
compensates for the velocity command value .omega.*.sub.(ave) by
using the velocity command-value compensation amount
.DELTA..omega.* generated by the load-pulsation compensating unit
20 and compensates for the carrier by using the carrier-frequency
command-value compensation amount .DELTA.fc* generated by the
load-pulsation compensating unit 20, without controlling the
response speed of the velocity control unit 24 and the current
control unit 25. Periodic pulsation in the load torque is thereby
suppressed.
The operation of the load-pulsation compensating unit 20 is
described below in detail. FIG. 7 is a diagram illustrating a
relation between the load torque .tau..sub.l, the actual rotor
rotation speed .omega..sub.m, the velocity command value .omega.*,
the velocity command-value compensation amount .DELTA..omega.*, and
the carrier-frequency command-value compensation amount .DELTA.fc*
in a case where a load in which toque pulsation occurs at a
constant period is connected to the motor 2. .tau..sub.l indicates
an instantaneous value of the load torque. .tau..sub.l(ave)
indicates the average value of the load torque. .omega.*.sub.(ave)
indicates the average rotation-speed command value that is a
velocity command value provided from a higher-level controller for
the motor control unit 18.
The velocity command-value compensation amount .DELTA..omega.* is
set on the basis of conditions represented in the following
Inequalities (2) and (3). if .tau..sub.l>.tau..sub.m then
.DELTA..omega.*>0 (2) if .tau..sub.l<.tau..sub.m then
.DELTA..omega.*<0 (3)
The conditions represented in Inequalities (2) and (3) are derived
as described below. When the load-pulsation compensating unit 20
does not function, that is, when ".omega.*=.omega.*.sub.(ave)",
then the actual motor rotation speed .omega..sub.m pulsates on the
basis of the load-torque pulsation. In particular, for the
condition represented in Inequality (2), the actual motor rotation
speed .omega..sub.m is reduced on the basis of Equation (1). For
the condition represented in Inequality (3), the actual motor
rotation speed .omega..sub.m is increased on the basis of Equation
(1).
The load-pulsation compensating unit 20 changes the velocity
command value .omega.* by the velocity command-value compensation
amount .DELTA..omega.*. Specifically, for the condition represented
in Inequality (2), the load-pulsation compensating unit 20
increases the velocity command value .omega.* so that the actual
motor rotation speed .omega..sub.m is not reduced. For the
condition represented in Inequality (3), the load-pulsation
compensating unit 20 reduces the velocity command value .omega.* so
that the actual motor rotation speed .omega..sub.m is not
increased. Thus, the power converting apparatus 1 can suppress
pulsation in the actual motor rotation speed .omega..sub.m through
the operation of the load-pulsation compensating unit 20.
Next, a procedure for determining the velocity command-value
compensation amount .DELTA..omega.* is described. In a case where
the load torque pulsates at a constant period, a pulsating
component of the load torque can be identified in advance. Thus,
the velocity command-value compensation amount .DELTA..omega.* can
be determined in advance at the time of designing a motor control
system and accordingly the feed-forward control can be used. It is
possible to have a configuration in which the velocity
command-value compensation amount .DELTA..omega.* based on the
load-torque pulsation is stored in a memory as a map. The inverter
control unit 13 may be implemented by a microcomputer or a Digital
Signal Processor (DSP) including the memory. The load-pulsation
compensating unit 20 reads the velocity command-value compensation
amount .DELTA..omega.* stored in the memory and compensates for the
velocity command value .omega.*.sub.(ave) by using the velocity
command-value compensation amount .DELTA..omega..
The load-pulsation compensating unit 20 according to the first
embodiment selects the velocity command-value compensation amount
.DELTA..omega.* on the basis of the velocity command value
.DELTA.*.sub.(ave) on the assumption that the load torque becomes
higher on the basis of the actual rotor rotation speed
.omega..sub.m as in a case of a compressor. The correlation between
the conditions represented in Inequalities (2) and (3) and a
procedure for selecting the velocity command-value compensation
amount .DELTA..omega.* will be described later in a second
embodiment.
Next, the operation of the synchronous PWM control unit 19 is
described. The carrier-frequency command-value compensation amount
.DELTA.fc* is set on the basis of the conditions represented in the
following Inequalities (4) and (5). if .tau..sub.l>.tau..sub.m
then .DELTA.fc*>0 (4) if .tau..sub.l<.tau..sub.m then
.DELTA.fc*<0 (5)
The conditions represented in Inequalities (4) and (5) are derived
as described below. As described above, on the basis of
Inequalities (2) and (3), the velocity command value .omega.* is
updated to reflect the velocity command-value compensation amount
.DELTA..omega.* based on the load-torque pulsation. Thus, the
frequency of a voltage output from the inverter 12 also needs to be
changed on the basis of the velocity command value .omega.*.
Accordingly, in order to maintain an integer multiple relation
between the carrier frequency and the frequency of the voltage
output from the inverter 12, a compensation amount also needs to be
set for the synchronous PWM control unit 19 on the basis of the
velocity command-value compensation amount .DELTA..omega.*.
A procedure for determining the carrier-frequency command-value
compensation amount .DELTA.fc* is described below. The synchronous
PWM control is to control the carrier frequency so as to become an
integer multiple of the frequency of the voltage output from the
inverter 12. Thus, the carrier-frequency command-value compensation
amount .DELTA.fc* can be calculated on the basis of the following
Equation (6). N in Equation (6) indicates the number of pulses in
the synchronous PWM control. Equation (6) is an example of
calculation of the carrier-frequency command-value compensation
amount .DELTA.fc*. .DELTA.fc*=N.times..DELTA..omega.* (6)
The correlation between the conditions represented in Inequalities
(4) and (5) and the procedure for selecting the carrier-frequency
command-value compensation amount .DELTA.fc* will be described
later in the second embodiment.
In a case where a load in which torque pulsation occurs at a
constant period is connected to the motor 2, the power converting
apparatus 1 according to the first embodiment compensates for the
velocity command value .omega.*.sub.(ave) by using the velocity
command-value compensation amount .DELTA..omega.* generated by the
load-pulsation compensating unit 20. Also, the power converting
apparatus 1 compensates for the carrier by using the
carrier-frequency command-value compensation amount .DELTA.fc*
generated by the load-pulsation compensating unit 20. Thus, even
when the load torque pulsates periodically, the power converting
apparatus 1 can still perform synchronous PWM modulation in a
stable manner.
Second Embodiment
The power converting apparatus 1 according to the first embodiment
selects the velocity command-value compensation amount
.DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc* in accordance with the relation
between the motor output torque .tau..sub.m and the load torque
.tau..sub.l by using the conditions represented in Inequalities (2)
to (5).
However, it may be difficult to detect the load torque during
driving of the motor 2 in some cases. The velocity command-value
compensation amount .DELTA..omega.* and the carrier-frequency
command-value compensation amount .DELTA.fc* are determined in
advance on the basis of the load torque, and thereafter stored as a
map in the memory of the inverter control unit 13. Even in that
case, if the load torque cannot be measured or detected during
operation, an alternative indicator is needed to select the
velocity command-value compensation amount .DELTA..omega.* and the
carrier-frequency command-value compensation amount .DELTA.fc*.
In the second embodiment, a case is described where the motor
rotation position .theta. is used as the alternative indicator.
FIG. 8 is a diagram illustrating a relation between the load torque
.tau..sub.l, the actual rotor rotation speed .omega..sub.m, and the
actual rotor rotation position .theta..sub.1 of the motor 2 in a
case where a load in which torque pulsation occurs at a constant
period is connected to the motor 2.
In FIG. 8, it is assumed that the load torque pulsates
periodically. Periodicity is correlated with the actual rotor
rotation position .theta..sub.1. FIG. 8 illustrates a case where
the load torque .tau..sub.l pulsates once per period of the actual
rotor rotation position .theta..sub.1. However, the load torque
.tau..sub.l may pulsate twice per period of the actual rotor
rotation position .theta..sub.1.
The load-pulsation compensating unit 20 can select the velocity
command-value compensation amount .DELTA..omega.* and the
carrier-frequency command-value compensation amount .DELTA.fc* on
the basis of the actual rotor rotation position .theta..sub.1.
In a case of driving with a position sensor, that is, a
configuration including a position detection mechanism, the actual
rotor rotation position .theta..sub.1 can be directly detected.
However, the power converting apparatus 1 according to the second
embodiment executes sensorless position control and thus does not
include a position detection mechanism or cannot directly detect
the actual rotor rotation position .theta..sub.1. Accordingly, the
power converting apparatus 1 according to the second embodiment
uses the rotor magnetic-pole position .theta. calculated by the
estimating unit 23.
FIG. 9 is a diagram illustrating a configuration of a motor control
unit 41 and a load-pulsation compensating unit 43 according to the
second embodiment. The motor control unit 41 and the load-pulsation
compensating unit 43 are different in configuration from the motor
control unit 18 and the load-pulsation compensating unit 20
according to the first embodiment in that the rotor magnetic-pole
position .theta. is input to a compensating unit 44 and the
velocity command-value compensation amount .DELTA..omega.* is
selected on the basis of the input rotor magnetic-pole position
.theta.. The other constituent elements are identical to those of
the motor control unit 18 and the load-pulsation compensating unit
20 according to the first embodiment, and therefore such
constituent elements are denoted by like reference signs and
explanations thereof are omitted.
An estimating unit 42 calculates the rotor magnetic-pole position
.theta. and the estimated velocity value .omega. of the motor 2 on
the basis of the electric currents Id and Iq and the voltage
command values Vd* and Vq* generated by the current control unit
25. The estimating unit 42 outputs the estimated velocity value
.omega. to the velocity control unit 24 and outputs the rotor
magnetic-pole position .theta. to the voltage-command calculating
unit 26 and the load-pulsation compensating unit 43.
The load-pulsation compensating unit 43 includes the compensating
unit 44 that calculates the velocity command-value compensation
amount .DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc*; and an adder 45 that generates the
velocity command value .omega.*.
On the basis of the rotor magnetic-pole position .theta., the
compensating unit 44 calculates the velocity command-value
compensation amount .DELTA..omega.* and the carrier-frequency
command-value compensation amount .DELTA.fc*. The adder 45 adds the
velocity command value .omega.*.sub.(ave) and the velocity
command-value compensation amount .DELTA..omega.* together to
generate the velocity command value .omega.*.
FIG. 10 is a diagram illustrating the configuration of the
synchronous PWM control unit 19 and the compensating unit 44
according to the second embodiment. The synchronous PWM control
unit 19 and the compensating unit 44 according to the second
embodiment are different in configuration from the synchronous PWM
control unit 19 and the compensating unit 31 according to the first
embodiment in that the rotor magnetic-pole position .theta. is
input to the compensating unit 44 and the carrier-frequency
command-value compensation amount .DELTA.fc* is selected on the
basis of the input rotor magnetic-pole position .theta.. The other
constituent elements are identical to those of the synchronous PWM
control unit 19 according to the first embodiment, and therefore
such constituent elements are denoted by like reference signs and
explanations thereof are omitted.
Thus, the power converting apparatus 1 according to the second
embodiment compensates for the velocity command value
.omega.*.sub.(ave) by using the velocity command-value compensation
amount .DELTA..omega.* selected on the basis of the rotor
magnetic-pole position of the motor 2. Also, the power converting
apparatus 1 compensates for the carrier by using the
carrier-frequency command-value compensation amount .DELTA.fc*
selected on the basis of the rotor magnetic-pole position of the
motor 2. Consequently, in a case where periodic pulsation has
occurred in a load connected to the motor 2, the frequency of a
synchronous PWM signal output from the synchronous PWM control unit
19 is periodically varied on the basis of the rotor magnetic-pole
position of the motor 2. Thus, even when the load torque pulsates
periodically, the power converting apparatus 1 can still perform
synchronous PWM modulation in a stable manner.
Third Embodiment
In the second embodiment, the motor rotation position .theta. is
defined as an indicator for selecting the velocity command-value
compensation amount .DELTA..omega.* and the carrier-frequency
command-value compensation amount .DELTA.fc*. In a third
embodiment, a case is described in which the q-axis current Iq is
defined as an indicator for selecting the velocity command-value
compensation amount .DELTA..omega.* and the carrier-frequency
command-value compensation amount .DELTA.fc*.
FIG. 11 is a diagram illustrating a relation between the load
torque .tau..sub.l, the actual rotor rotation speed .omega..sub.m,
and the q-axis current Iq. Iq.sub.(ave) indicates the average
motor-current value.
The q-axis current Iq that flows when the load torque T.sub.1 is
higher than the average load-torque value .tau..sub.l(ave), that
is, when ".tau..sub.l>.tau..sub.l(ave)" is greater than the
average motor-current value Iq.sub.(ave) that flows when
".tau..sub.l=.tau..sub.l(ave)". In contrast, the q-axis current Iq
that flows when the load torque .tau..sub.l is lower than the
average load-torque value .tau..sub.l(ave), that is, when
".tau..sub.l<.tau..sub.l(ave)" is less than the average
motor-current value Iq.sub.(ave) that flows when
".tau..sub.l=.tau..sub.l(ave)".
Thus, the state of the load torque can be identified on the basis
of the magnitude of the q-axis current Iq. Accordingly, the q-axis
current Iq can be defined as an indicator for selecting the
velocity command-value compensation amount .DELTA..omega.* and the
carrier-frequency command-value compensation amount .DELTA.fc*. The
q-axis current Iq indicates a torque component.
FIG. 12 is a diagram illustrating a configuration of a motor
control unit 51 and a load-pulsation compensating unit 53 according
to the third embodiment. The motor control unit 51 and the
load-pulsation compensating unit 53 are different in configuration
from the motor control unit 18 and the load-pulsation compensating
unit 20 according to the first embodiment in that the q-axis
current Iq is input to a compensating unit 54 and the velocity
command-value compensation amount .DELTA..omega.* is selected on
the basis of the input q-axis current Iq. The other constituent
elements are identical to those of the motor control unit 18 and
the load-pulsation compensating unit 20 according to the first
embodiment, and therefore such constituent elements are denoted by
like reference signs and explanations thereof are omitted.
On the basis of the rotor magnetic-pole position .theta. of the
motor 2, a converting unit 52 converts the phase currents Iu, Iv,
and Iw, which are three phase currents, into two phase currents,
and converts the two phase currents into the d-axis current Id and
the q-axis current Iq on the d-q coordinate axes. The converting
unit 52 outputs the d-axis current Id to the estimating unit 23 and
the current control unit 25, and outputs the q-axis current Iq to
the estimating unit 23, the current control unit 25, and the
load-pulsation compensating unit 53.
The load-pulsation compensating unit 53 includes the compensating
unit 54 that calculates the velocity command-value compensation
amount .DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc*; and an adder 55 that generates the
velocity command value .omega.*.
On the basis of the q-axis current Iq, the compensating unit 54
calculates the velocity command-value compensation amount
.DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc*. The adder 55 adds the velocity
command value .omega.*.sub.(ave) and the velocity command-value
compensation amount .DELTA..omega.* together to generate the
velocity command value .omega.*.
FIG. 13 is a diagram illustrating the configuration of the
synchronous PWM control unit 19 and the compensating unit 54
according to the third embodiment. The synchronous PWM control unit
19 and the compensating unit 54 according to the third embodiment
are different in configuration from the synchronous PWM control
unit 19 and the compensating unit 31 according to the first
embodiment in that the q-axis current Iq is input to the
compensating unit 54 and the carrier-frequency command-value
compensation amount .DELTA.fc* is selected on the basis of the
input q-axis current Iq. The other constituent elements are
identical to those of the synchronous PWM control unit 19 according
to the first embodiment, and therefore such constituent elements
are denoted by like reference signs and explanations thereof are
omitted.
Thus, the power converting apparatus 1 according to the third
embodiment compensates for the velocity command value
.omega.*.sub.(ave) by using the velocity command-value compensation
amount .DELTA..omega.* selected on the basis of the q-axis current
Iq. Also, the power converting apparatus 1 compensates for the
carrier by using the carrier-frequency command-value compensation
amount .DELTA.fc* selected on the basis of the q-axis current Iq.
Consequently, in a case where periodic pulsation has occurred in a
load connected to the motor 2, the frequency of a synchronous PWM
signal output from the synchronous PWM control unit 19 is
periodically varied on the basis of the q-axis current Iq that is
an electric current flowing to the motor 2. Thus, even when the
load torque pulsates periodically, the power converting apparatus 1
can still perform synchronous PWM modulation in a stable
manner.
A description has been given where the power converting apparatus 1
in the second embodiment is configured such that the velocity
command-value compensation amount .DELTA..omega.* and the
carrier-frequency command-value compensation amount .DELTA.fc* are
selected on the basis of the rotor magnetic-pole position and the
power converting apparatus 1 in the third embodiment is configured
such that the velocity command-value compensation amount
.DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc* are selected on the basis of the
q-axis current Iq. It is also permissible to employ a configuration
to select the velocity command-value compensation amount
.DELTA..omega.* and the carrier-frequency command-value
compensation amount .DELTA.fc* on the basis of the maximum value of
the phase currents Iu, Iv, and Iw, the peak-to-peak value of the
phase currents Iu, Iv, and Iw, the effective value of the phase
currents Iu, Iv, and Iw, the average value of the phase currents
Iu, Iv, and Iw, or any of the phase currents Iu, Iv, and Iw.
Fourth Embodiment
The power converting apparatus 1 according to the first to third
embodiments may be included in a heat pump device 100. FIG. 14 is a
diagram illustrating the configuration of the heat pump device 100
according to a fourth embodiment.
The heat pump device 100 includes a refrigeration cycle in which a
compressor 101 having a compression mechanism that compresses a
refrigerant, a four-way valve 102 that changes the direction of
refrigerant gas, heat exchangers 103 and 104, and an expansion
mechanism 105 are sequentially connected via a refrigerant pipe
106. The direction of refrigerant gas is switched to a first
direction by the four-way valve 102, and thereby the heat exchanger
103 serves as an evaporator while the heat exchanger 104 serves as
a condenser. In addition, the direction of the refrigerant gas is
switched to a second direction by the four-way valve 102, and
thereby the heat exchanger 103 serves as a condenser while the heat
exchanger 104 serves as an evaporator. In FIG. 14, the four-way
valve 102 has switched the direction of refrigerant gas to the
first direction.
The compressor 101 includes a compression mechanism 107 that
compresses a refrigerant and the motor 2 that operates the
compression mechanism 107. The motor 2 is a three-phase motor
including windings for three phases that are a U-phase, a V-phase,
and a W-phase. The motor 2 is driven by being supplied with an
alternating-current voltage from the power converting apparatus
1.
In a case where the compression mechanism 107 in which torque
pulsation occurs at a constant period is connected to the motor 2,
the heat pump device 100 according to the fourth embodiment
compensates for the velocity command value .omega.*.sub.(ave) by
using the velocity command-value compensation amount
.DELTA..omega.* and also compensates for the carrier by using the
carrier-frequency command-value compensation amount .DELTA.fc*.
Thus, even when the load torque pulsates periodically, the heat
pump device 100 can still perform synchronous PWM modulation in a
stable manner. The heat pump device 100 is applicable to an air
conditioning device.
As illustrated in FIG. 15, the inverter control unit 13 in the
power converting apparatus 1 according to the first to third
embodiments may be configured from a CPU 201 that performs
calculation, a memory 202 that stores therein a program read by the
CPU 201, and an interface 203 that inputs and outputs signals.
Specifically, a program that executes functions of the inverter
control unit 13 is stored in the memory 202. The voltage Vdc
detected by the direct-current-voltage detecting unit 14 and the
electric current Idc detected by the current detecting unit 15 are
input to the CPU 201 via the interface 203. The CPU 201 then
generates the synchronous PWM signals UP, VP, WP, UN, VN, and WN
and outputs the generated synchronous PWM signals UP, VP, WP, UN,
VN, and WN via the interface 203. The synchronous PWM signals UP,
VP, WP, UN, VN, and WN output from the interface 203 are applied to
the inverter 12.
The configurations described in the above embodiments are only
examples of the content of the present invention. The
configurations can be combined with other well-known techniques,
and part of each of the configurations can be omitted or modified
without departing from the scope of the present invention.
* * * * *