U.S. patent number 10,153,872 [Application Number 15/705,115] was granted by the patent office on 2018-12-11 for apparatus for transmitting and receiving a signal and method of transmitting and receiving a signal.
This patent grant is currently assigned to LG ELECTRONICS INC.. The grantee listed for this patent is LG ELECTRONICS INC.. Invention is credited to Woo Suk Ko, Sang Chul Moon.
United States Patent |
10,153,872 |
Ko , et al. |
December 11, 2018 |
Apparatus for transmitting and receiving a signal and method of
transmitting and receiving a signal
Abstract
The present invention relates to method of transmitting and
receiving signals and a corresponding apparatus. One aspect of the
present invention relates to a method of receiving a signal, which
includes an L1 signaling region where the L1 signaling has an
adaptive L1 block structure for increased spectrum efficiency in a
channel bonding environment.
Inventors: |
Ko; Woo Suk (Seoul,
KR), Moon; Sang Chul (Seoul, KR) |
Applicant: |
Name |
City |
State |
Country |
Type |
LG ELECTRONICS INC. |
Seoul |
N/A |
KR |
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Assignee: |
LG ELECTRONICS INC. (Seoul,
KR)
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Family
ID: |
41119726 |
Appl.
No.: |
15/705,115 |
Filed: |
September 14, 2017 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20180006777 A1 |
Jan 4, 2018 |
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Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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15391713 |
Dec 27, 2016 |
9806861 |
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12922681 |
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PCT/KR2009/002504 |
May 12, 2009 |
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61109944 |
Oct 31, 2008 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H04L
5/0007 (20130101); H05K 999/99 (20130101); H04L
5/0053 (20130101); H04L 27/34 (20130101); H04L
2001/0093 (20130101); H04L 69/323 (20130101); H04L
1/0072 (20130101) |
Current International
Class: |
H04L
27/34 (20060101); H03M 13/25 (20060101); H04L
12/18 (20060101); H04L 5/00 (20060101); H04L
1/00 (20060101); H04L 29/08 (20060101) |
Field of
Search: |
;725/109 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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1473423 |
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Feb 2004 |
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CN |
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101167322 |
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Apr 2008 |
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CN |
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101198179 |
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Jun 2008 |
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CN |
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101202606 |
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Jun 2008 |
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CN |
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101212392 |
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Jul 2008 |
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CN |
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1976317 |
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Oct 2008 |
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EP |
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1983678 |
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Oct 2008 |
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EP |
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2034758 |
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Mar 2009 |
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EP |
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2007-148585 |
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Dec 2007 |
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WO |
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Other References
The State Intellectual Property Office of the People's Republic of
China Application No. 201410408849.2, Office Action dated Nov. 17,
2016, 12 pages. cited by applicant .
Digital Video Broadcasting (DVB), "Frame Structure Channel Coding
and Modulation for a Second Generation Digital Terrestrial
Television Broadcasting System (DVB-T2)," DVB Document A122, Jun.
2008. cited by applicant .
European Telecommunications Standards Institute (ETSI), "Digital
Video Broadcasting (DVB), Second Generation Framing Structure,
Channel Coding and Modulation Systems for Broadcasting, Interactive
Services, News Gathering and Other Broadband Satellite
Applications," ETSI EN 302 307 V1.1.2, Jun. 2006. cited by
applicant .
Sony, "Response to the DVB-C2 Call for Technologies (CfT)," Jun.
2008, 57 pages. cited by applicant .
The State Intellectual Property Office of the People's Republic of
China Application Serial No. 200980129244.7, Office Action dated
Apr. 9, 2013, 8 pages. cited by applicant .
ETSI,"Digital Video Broadcasting (DVB); Frame structure channel
coding and modulation for a second generation digital terrestrial
television broadcasting system (DVB-T2)", Draft DVB-T2
Specification Version 0.5.5, Mar. 2008, XP017802195. cited by
applicant .
Sony, "Response to the DVB-C2 Call for Technologies (CfT)", Jun.
2008, XP017830432. cited by applicant .
LG Electronics Inc, "L1 signaling part2", Jan. 2008, XP017830550.
cited by applicant .
ETSI, "Digital Video Broadcasting (DVB); Frame structure channel
coding and modulation for a second generation transmission system
for cable systems (DVB-C2)", Draft ETSI EN 302 769 V0.9, Mar. 2009,
XP017800109. cited by applicant.
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Primary Examiner: Nawaz; Asad
Assistant Examiner: Harley; Jason
Attorney, Agent or Firm: Lee, Hong, Degerman, Kang &
Waimey
Parent Case Text
CROSS-REFERENCE TO RELATED APPLICATIONS
This application is a continuation of U.S. application Ser. No.
15/391,713, filed Dec. 27, 2016, now U.S. Pat. No. 9,806,861, which
is a continuation of U.S. application Ser. No. 12/922,681, filed
Sep. 14, 2010, now abandoned, which is a 371 U.S. national stage
application of International Application No. PCT/KR2009/002504,
filed on May 12, 2009, which claims priority to U.S. Provisional
Application Ser. No. 61/109,944, filed on Oct. 31, 2008, the
contents of all of which are incorporated by reference herein in
their entireties.
Claims
The invention claimed is:
1. A method of transmitting broadcast data in a broadcast
transmitting system, the method comprising: Cyclic Redundancy Check
(CRC) encoding Physical Layer Pipe (PLP) data of a PLP; inner
encoding the CRC-encoded PLP data; interleaving the inner encoded
PLP data; mapping the interleaved PLP data onto constellations
using a non-uniform constellation M-Quadrature Amplitude Modulation
(QAM) scheme, wherein M is 64, 256,1024, or 4096; outer encoding
Layer 1 (L1) signaling information for signaling the PLP; padding
the outer encoded L1 signaling information with at least one zero
bit; inner encoding the outer encoded L1 signaling information and
the at least one zero bit, resulting in inner encoded L1 signaling
information; removing the at least one zero bit from the inner
encoded L1 signaling information; interleaving the inner encoded L1
signaling information from which the at least one zero bit is
removed; mapping the interleaved L1 signaling information onto
constellations using a non-uniform constellation N-QAM scheme,
wherein N is an integer greater than 1; building a signal frame
including the mapped PLP data and the mapped L1 signaling
information; modulating the signal frame by an Orthogonal Frequency
Division Multiplexing (OFDM) method; and transmitting a
broadcasting signal including data of the modulated signal
frame.
2. The method of claim 1, wherein the L1 signaling information
includes modulation information that identifies the non-uniform
constellation M-QAM.
3. The method of claim 1, further comprising puncturing at least
one parity bit generated by inner encoding bits of the outer
encoded L1 signaling information and the at least one zero bit.
4. The method of claim 1, wherein the L1 signaling information
includes PLP identification (ID) information that identifies the
PLP, PLP start information that identifies a start position of the
PLP, and PLP TYPE information that indicates a type of the PLP.
5. A broadcast transmitting system for transmitting broadcast data,
the broadcast transmitting system comprising: a first encoder to
Cyclic Redundancy Check (CRC) encode Physical Layer Pipe (PLP) data
of a PLP; a second encoder to inner encode the CRC-encoded PLP
data; a first interleaver to interleave the inner encoded PLP data;
a first mapper to map the interleaved PLP data onto constellations
using a non-uniform constellation M-QAM scheme, wherein M is 64,
256, 1024,or 4096; a third encoder to outer encode L1 signaling
information for signaling the PLP; a zero padding circuit to pad
the outer encoded L1 signaling information with at least one zero
bit; a fourth encoder to inner encode the outer encoded L1
signaling information and the at least one zero bit, resulting in
inner encoded L1 signaling information; a zero removal circuit to
remove the at least one zero bit from the inner encoded L1
signaling information; a second interleaver to interleave the inner
encoded L1 signaling information from which the at least one zero
bit is removed; a second mapper to map the interleaved L1 signaling
information onto constellations using a non-uniform constellation
N-QAM scheme, wherein N is an integer greater than 1; a frame
builder to build a signal frame including the mapped PLP data and
the mapped L1 signaling information; a modulator to modulate the
signal frame by an Orthogonal Frequency Division Multiplexing
(OFDM) method; and a transmitter to transmit a broadcasting signal
including data of the modulated signal frame.
6. The broadcast transmitting system of claim 5, wherein the L1
signaling information includes modulation information that
identifies the non-uniform constellation M-QAM.
7. The broadcast transmitting system of claim 5, further comprising
a parity puncturing circuit to puncture at least one parity bit
generated by inner encoding bits of the outer encoded L1 signaling
information and the at least one zero bit.
8. The broadcast transmitting system of claim 5, wherein the L1
signaling information includes PLP ID information that identifies
the PLP, PLP start information that identifies a start position of
the PLP, and PLP TYPE information that indicates a type of the
PLP.
9. A method of receiving broadcast data in a broadcast receiving
system, the method comprising: receiving a broadcast signal
including a signal frame that contains Physical Layer Pipe (PLP)
data of a PLP and L1 signaling information for signaling the PLP;
demodulating the received broadcast signal by an Orthogonal
Frequency Division Multiplexing (OFDM) method; parsing the signal
frame in the demodulated broadcast signal; demapping the PLP data
in the signal frame using a non-uniform constellation M-QAM scheme,
wherein M is 64, 256, 1024,or 4096; deinterleaving the demapped PLP
data; inner decoding the deinterleaved PLP data; Cyclic Redundancy
Check (CRC) decoding the inner decoded PLP data; demapping the L1
signaling information in the signal frame using a non-uniform
constellation N-QAM scheme, wherein N is an integer greater than 1;
deinterleaving the demapped L1 signaling information; padding the
deinterleaved L1 signaling information with at least one zero bit;
inner decoding the deinterleaved L1 signaling information and the
at least one zero bit, resulting inner decoded L1 signaling
information; removing the at least one zero bit from the inner
decoded L1 signaling information; and outer decoding the inner
decoded L1 signaling information from which the at least one zero
bit is removed.
10. The method of claim 9, wherein the L1 signaling information
includes modulation information that identifies the non-uniform
constellation M-QAM.
11. The method of claim 9, further comprising depuncturing parity
bits in the deinterleaved L1 signaling information.
12. The method of claim 9, wherein the L1 signaling information
includes PLP ID information that identifies the PLP, PLP start
information that identifies a start position of the PLP, and PLP
TYPE information that indicates a type of the PLP.
13. A broadcast receiving system for receiving broadcast data, the
broadcast receiving system comprising: a tuner to receive a
broadcast signal including a signal frame that contains Physical
Layer Pipe (PLP) data of a PLP and L1 signaling information for
signaling the PLP; a demodulator to demodulate the received
broadcast signal by an Orthogonal Frequency Division Multiplexing
(OFDM) method; a frame parser to parse the signal frame in the
demodulated broadcast signal; a first demapper to demap the PLP
data in the signal frame using a non-uniform constellation M-QAM
scheme, wherein M is 64, 256, 1024,or 4096; a first deinterleaver
to deinterleave the demapped PLP data; a first decoder to inner
decode the deinterleaved PLP data; a second decoder to CRC decode
the inner decoded PLP data; a second damapper to demap the L1
signaling information in the signal frame using a non-uniform
constellation N-QAM scheme, wherein N is an integer greater than 1;
a second deinterleaver to deinterleave the demapped L1 signaling
information; a zero padding circuit to pad the deinterleaved L1
signaling information with at least one zero bit; a third decoder
to inner decode the deinterleaved L1 signaling information and the
at least one zero bit, resulting inner decoded L1 signaling
information; a zero removal circuit to remove the at least one zero
bit from the inner decoded L1 signaling information; and a fourth
decoder to outer decode the inner decoded L1 signaling information
from which the at least one zero bit is removed.
14. The broadcast receiving system of claim 13, wherein the L1
signaling information includes modulation information that
identifies the non-uniform constellation M-QAM.
15. The broadcast receiving system of claim 13, comprising a parity
depuncturing circuit to depuncture parity bits in the deinterleaved
L1 signaling information.
16. The broadcast receiving system of claim 13, wherein the L1
signaling information includes PLP ID information that identifies
the PLP, PLP start information that identifies a start position of
the PLP, and PLP TYPE information that indicates a type of the PLP.
Description
TECHNICAL FIELD
The present invention relates to a method for transmitting and
receiving a signal and an apparatus for transmitting and receiving
a signal, and more particularly, to a method for transmitting and
receiving a signal and an apparatus for transmitting and receiving
a signal, which are capable of improving data transmission
efficiency.
BACKGROUND ART
As a digital broadcasting technology has been developed, users have
received a high definition (HD) moving image. With continuous
development of a compression algorithm and high performance of
hardware, a better environment will be provided to the users in the
future. A digital television (DTV) system can receive a digital
broadcasting signal and provide a variety of supplementary services
to users as well as a video signal and an audio signal.
Digital Video Broadcasting (DVB)-C2 is the third specification to
join DVB's family of second generation transmission systems.
Developed in 1994, today DVB-C is deployed in more than 50 million
cable tuners worldwide. In line with the other DVB second
generation systems, DVB-C2 uses a combination of Low-density
parity-check (LDPC) and BCH codes. This powerful Forward Error
correction (FEC) provides about 5 dB improvement of
carrier-to-noise ratio over DVB-C. Appropriate bit-interleaving
schemes optimize the overall robustness of the FEC system. Extended
by a header, these frames are called Physical Layer Pipes (PLP).
One or more of these PLPs are multiplexed into a data slice. Two
dimensional interleaving (in the time and frequency domains) is
applied to each slice enabling the receiver to eliminate the impact
of burst impairments and frequency selective interference such as
single frequency ingress.
DISCLOSURE OF INVENTION
Technical Problem
With the development of these digital broadcasting technologies, a
requirement for a service such as a video signal and an audio
signal increased and the size of data desired by users or the
number of broadcasting channels gradually increased.
Technical Solution
Accordingly, the present invention is directed to a method for
transmitting and receiving a signal and an apparatus for
transmitting and receiving a signal that sub-stantially obviate one
or more problems due to limitations and disadvantages of the
related art.
An object of the present invention is to provide a method of
transmitting at least one broadcasting signal frame having PLP
(Physical Layer Pipe) data and preamble data, the method
comprising: mapping bits of the PLP data into PLP data symbols and
bits of the preamble data into preamble data symbols; building at
least one data slice based on the PLP data symbols; building a
signal frame based on the preamble data symbols and the data
slices, the preamble data symbols having L1 signaling information
for signaling the data slices, modulating the signal frame by an
Orthogonal Frequency Division Multiplexing (OFDM) method; and
transmitting the modulated signal frame, wherein the L1 signaling
information has data slice ID information that identifies the data
slice.
Another aspect of the present invention provides a method of
receiving broadcasting signal, comprising: demodulating the
received signal by use of an Orthogonal Frequency Division
Multiplexing (OFDM) method; obtaining a signal frame from the
demodulated signals, the signal frame comprising preamble symbols
and PLP (Physical Layer Pipe) data symbols, the preamble symbols
having L1 signaling information, the L1 signaling information
having data slice ID information that identifies the data slice,
the data slice being identical with a group of data symbols;
demapping into bits for the preamble symbols and bits for the PLP
data symbols; and decoding the bits for the preamble symbols by a
shortened and punctured LDPC (low density parity check) decoding
scheme.
Yet another aspect of the present invention provides a transmitter
of transmitting at least one broadcasting signal frame having PLP
(Physical Layer Pipe) data and preamble data, comprising: a mapper
configured to map bits of the PLP data into PLP data symbols and
bits of the preamble data into preamble data symbols; a data slice
builder configured to build at least one data slice based on the
PLP data symbols; a frame builder configured to build a signal
frame based on the preamble data symbols and the data slices, the
preamble data symbols having L1 signaling information for signaling
the data slices, the L1 signaling information having data slice ID
information that identifies the data slice; a modulator configured
to modulate the signal frame by an Orthogonal Frequency Division
Multiplexing (OFDM) method; and a transmission unit configured to
transmit the modulated signal frame.
Yet another aspect of the present invention provides a receiver of
receiving broadcasting signal, comprising; a demodulator configured
to demodulate the received signal by use of an Orthogonal Frequency
Division Multiplexing (OFDM) method; a frame Parser configured to
obtain a signal frame from the demodulated signals, the signal
frame comprising preamble symbols and PLP (Physical Layer Pipe)
data symbols, the preamble symbols having L1 signaling information,
the L1 signaling information having data slice ID information that
identifies the data slice, the data slice being identical with a
group of data symbols; a demapper configured to demap the preamble
symbols into preamble bits and PLP data symbols into PLP data bits;
and a decoder configured to decode the preamble bits by a shortened
and a punctured LDPC (low density parity check) decoding
scheme.
BRIEF DESCRIPTION OF DRAWINGS
The accompanying drawings, which are included to provide a further
understanding of the invention and are incorporated in and
constitute a part of this application, illustrate embodiment(s) of
the invention and together with the description serve to explain
the principle of the invention. In the drawings:
FIG. 1 is an example of 64-Quadrature amplitude modulation (QAM)
used in European DVB-T.
FIG. 2 is a method of Binary Reflected Gray Code (BRGC).
FIG. 3 is an output close to Gaussian by modifying 64-QAM used in
DVB-T.
FIG. 4 is Hamming distance between Reflected pair in BRGC.
FIG. 5 is characteristics in QAM where Reflected pair exists for
each I axis and Q axis.
FIG. 6 is a method of modifying QAM using Reflected pair of
BRGC.
FIG. 7 is an example of modified 64/256/1024/4096-QAM.
FIGS. 8-9 are an example of modified 64-QAM using Reflected Pair of
BRGC.
FIGS. 10-11 are an example of modified 256-QAM using Reflected Pair
of BRGC.
FIGS. 12-13 are an example of modified 1024-QAM using Reflected
Pair of BRGC(0.about.511).
FIGS. 14-15 are an example of modified 1024-QAM using Reflected
Pair of BRGC(512.about.1023).
FIGS. 16-17 are an example of modified 4096-QAM using Reflected
Pair of BRGC(0.about.511).
FIGS. 18-19 are an example of modified 4096-QAM using Reflected
Pair of BRGC(512.about.1023).
FIGS. 20-21 are an example of modified 4096-QAM using Reflected
Pair of BRGC(1024.about.1535).
FIGS. 22-23 are an example of modified 4096-QAM using Reflected
Pair of BRGC(1536.about.2047).
FIGS. 24-25 are an example of modified 4096-QAM using Reflected
Pair of BRGC(2048.about.2559).
FIGS. 26-27 are an example of modified 4096-QAM using Reflected
Pair of BRGC(2560.about.3071).
FIGS. 28-29 are an example of modified 4096-QAM using Reflected
Pair of BRGC(3072.about.3583).
FIGS. 30-31 are an example of modified 4096-QAM using Reflected
Pair of BRGC(3584.about.4095).
FIG. 32 is an example of Bit mapping of Modified-QAM where 256-QAM
is modified using BRGC.
FIG. 33 is an example of transformation of MQAM into Non-uniform
constellation.
FIG. 34 is an example of digital transmission system.
FIG. 35 is an example of an input processor.
FIG. 36 is an information that can be included in Base band
(BB).
FIG. 37 is an example of BICM.
FIG. 38 is an example of shortened/punctured encoder.
FIG. 39 is an example of applying various constellations.
FIG. 40 is another example of cases where compatibility between
conventional systems is considered.
FIG. 41 is a frame structure which comprises preamble for L1
signaling and data symbol for PLP data.
FIG. 42 is an example of frame builder.
FIG. 43 is an example of pilot insert (404) shown in FIG. 4.
FIG. 44 is a structure of SP.
FIG. 45 is a new SP structure or Pilot Pattern (PP) 5.
FIG. 46 is a suggested PP5' structure.
FIG. 47 is a relationship between data symbol and preamble.
FIG. 48 is another relationship between data symbol and
preamble.
FIG. 49 is an example of cable channel delay profile.
FIG. 50 is scattered pilot structure that uses z=56 and z=112.
FIG. 51 is an example of modulator based on OFDM.
FIG. 52 is an example of preamble structure.
FIG. 53 is an example of Preamble decoding.
FIG. 54 is a process for designing more optimized preamble.
FIG. 55 is another example of preamble structure
FIG. 56 is another example of Preamble decoding.
FIG. 57 is an example of Preamble structure.
FIG. 58 is an example of L1 decoding.
FIG. 59 is an example of analog processor.
FIG. 60 is an example of digital receiver system.
FIG. 61 is an example of analog processor used at receiver.
FIG. 62 is an example of demodulator.
FIG. 63 is an example of frame parser.
FIG. 64 is an example of BICM demodulator.
FIG. 65 is an example of LDPC decoding using
shortening/puncturing.
FIG. 66 is an example of output processor.
FIG. 67 is an example of L1 block repetition rate of 8 MHz.
FIG. 68 is an example of L1 block repetition rate of 8 MHz.
FIG. 69 is a new L1 block repetition rate of 7.61 MHz.
FIG. 70 is an example of L1 signaling which is transmitted in frame
header.
FIG. 71 is preamble and L1 Structure simulation result.
FIG. 72 is an example of symbol interleaver.
FIG. 73 is an example of an L1 block transmission.
BEST MODE FOR CARRYING OUT THE INVENTION
Reference will now be made in detail to the preferred embodiments
of the present invention, examples of which are illustrated in the
accompanying drawings. Wherever possible, the same reference
numbers will be used throughout the drawings to refer to the same
or like parts.
In the following description, the term "service" is indicative of
either broadcast contents which can be transmitted/received by the
signal transmission/reception apparatus.
Quadrature amplitude modulation (QAM) using Binary Reflected Gray
Code (BRGC) is used as modulation in a broadcasting transmission
environment where conventional Bit Interleaved Coded Modulation
(BICM) is used. FIG. 1 shows an example of 64-QAM used in European
DVB-T.
BRGC can be made using the method shown in FIG. 2. An n bit BRGC
can be made by adding a reverse code of (n-1) bit BRGC (i.e.,
reflected code) to a back of (n-1) bit, by adding 0s to a front of
original (n-1) bit BRGC, and by adding 1s to a front of reflected
code. The BRGC code made by this method has a Hamming distance
between adjacent codes of one (1). In addition, when BRGC is
applied to QAM, the Hamming distance between a point and the four
points which are most closely adjacent to the point, is one (1) and
the Hamming distance between the point and another four points
which are second most closely adjacent to the point, is two (2).
Such characteristics of Hamming distances between a specific
constellation point and other adjacent points can be dubbed as Gray
mapping rule in QAM.
To make a system robust against Additive White Gaussian Noise
(AWGN), distribution of signals transmitted from a transmitter can
be made close to Gaussian distribution. To be able to do that,
locations of points in constellation can be modified. FIG. 3 shows
an output close to Gaussian by modifying 64-QAM used in DVB-T. Such
constellation can be dubbed as Non-uniform QAM (NU-QAM).
To make a constellation of Non-uniform QAM, Gaussian Cumulative
Distribution Function (CDF) can be used. In case of 64, 256, or
1024 QAM, i.e., 2^N AMs, QAM can be divided into two independent
N-PAM. By dividing Gaussian CDF into N sections of identical
probability and by allowing a signal point in each section to
represent the section, a constellation having Gaussian distribution
can be made. In other words, coordinate xj of newly defined
non-uniform N-PAM can be defined as follows:
.intg..infin..times..times..pi..times..times..di-elect
cons..times..times..times..times..times..times. ##EQU00001##
FIG. 3 is an example of transforming 64QAM of DVB-T into NU-64QAM
using the above methods. FIG. 3 represents a result of modifying
coordinates of each I axis and Q axis using the above methods and
mapping the previous constellation points to newly defined
coordinates. In case of 32, 128, or 512 QAM, i.e., cross QAM, which
is not 2^N QAM, by modifying Pj appropriately, a new coordinate can
be found.
One embodiment of the present invention can modify QAM using BRGC
by using characteristics of BRGC. As shown in FIG. 4, the Hamming
distance between Reflected pair in BRGC is one because it differs
only in one bit which is added to the front of each code. FIG. 5
shows the characteristics in QAM where Reflected pair exists for
each I axis and Q axis. In this figure, Reflected pair exists on
each side of the dotted black line.
By using Reflected pairs existing in QAM, an average power of a QAM
constellation can be lowered while keeping Gray mapping rule in
QAM. In other words, in a constellation where an average power is
normalized as 1, the minimum Euclidean distance in the
constellation can be increased. When this modified QAM is applied
to broadcasting or communication systems, it is possible to
implement either a more noise-robust system using the same energy
as a conventional system or a system with the same performance as a
conventional system but which uses less energy.
FIG. 6 shows a method of modifying QAM using Reflected pair of
BRGC. FIG. 6a shows a constellation and FIG. 6b shows a flowchart
for modifying QAM using Reflected pair of BRGC. First, a target
point which has the highest power among constellation points needs
to be found. Candidate points are points where that target point
can move and are the closest neighbor points of the target point's
reflected pair. Then, an empty point (i.e., a point which is not
yet taken by other points) having the smallest power needs to be
found among the candidate points and the power of the target point
and the power of a candidate point are compared. If the power of
the candidate point is smaller, the target point moves to the
candidate point. These processes are repeated until an average
power of the points on constellation reaches a minimum while
keeping Gray mapping rule.
FIG. 7 shows an example of modified 64/256/1024/4096-QAM. The Gray
mapped values correspond to FIGS. 8.about.31 respectively. In
addition to these examples, other types of modified QAM which
enables identical power optimization can be realized. This is
because a target point can move to multiple candidate points. The
suggested modified QAM can be applied to, not only the
64/256/1024/4096-QAM, but also cross QAM, a bigger size QAM, or
modulations using other BRGC other than QAM.
FIG. 32 shows an example of Bit mapping of Modified-QAM where
256-QAM is modified using BRGC. FIG. 32a and FIG. 32b show mapping
of Most Significant Bits (MSB). Points designated as filled circles
represent mappings of ones and points designated as blank circles
represent mappings of zeros. In a same manner, each bit is mapped
as shown in figures from (a) through (h) in FIG. 32, until Least
Significant Bits (LSB) are mapped. As shown in FIG. 32,
Modified-QAM can enable bit decision using only I or Q axes as
conventional QAM, except for a bit which is next to MSB (FIG. 32c
and FIG. 32d). By using these characteristics, a simple receiver
can be made by partially modifying a receiver for QAM. An efficient
receiver can be implemented by checking both I and Q values only
when determining bit next to MSB and by calculating only I or Q for
the rest of bits. This method can be applied to Approximate LLR,
Exact LLR, or Hard decision.
By using the Modified-QAM or MQAM, which uses the characteristics
of above BRGC, Non-uniform constellation or NU-MQAM can be made. In
the above equation where Gaussian CDF is used, Pj can be modified
to fit MQAM. Just like QAM, in MQAM, two PAMs having I axis and Q
axis can be considered. However, unlike QAM where a number of
points corresponding to a value of each PAM axis are identical, the
number of points changes in MQAM. If a number of points that
corresponds to jth value of PAM is defined as nj in a MQAM where a
total of M constellation points exist, then Pj can be defined as
follows:
.intg..infin..times..times..pi..times..times..times..times..times..times.-
.times. ##EQU00002##
By using the newly defined Pj, MQAM can be transformed into
Non-uniform constellation. Pj can be defined as follows for the
example of 256-MQAM.
.di-elect cons. ##EQU00003##
FIG. 33 is an example of transformation of MQAM into Non-uniform
constellation. The NU-MQAM made using these methods can retain
characteristics of MQAM receivers with modified coordinates of each
PAM. Thus, an efficient receiver can be implemented. In addition, a
more noise-robust system than the previous NU-QAM can be
implemented. For a more efficient broadcasting transmission system,
hybridizing MQAM and NU-MQAM is possible. In other words, a more
noise-robust system can be implemented by using MQAM for an
environment where an error correction code with high code rate is
used and by using NU-MQAM otherwise. For such a case, a transmitter
can let a receiver have information of code rate of an error
correction code currently used and a kind of modulation currently
used such that the receiver can demodulate according to the
modulation currently used.
FIG. 34 shows an example of digital transmission system. Inputs can
comprise a number of MPEG-TS streams or GSE (General Stream
Encapsulation) streams. An input processor module 101 can add
transmission parameters to input stream and perform scheduling for
a BICM module 102. The BICM module 102 can add redundancy and
interleave data for transmission channel error correction. A frame
builder 103 can build frames by adding physical layer signaling
information and pilots. A modulator 104 can perform modulation on
input symbols in efficient methods. An analog processor 105 can
perform various processes for converting input digital signals into
output analog signals.
FIG. 35 shows an example of an input processor. Input MPEG-TS or
GSE stream can be transformed by input preprocessor into a total of
n streams which will be independently processed. Each of those
streams can be either a complete TS frame which includes multiple
service components or a minimum TS frame which includes service
component (i.e., video or audio). In addition, each of those
streams can be a GSE stream which transmits either multiple
services or a single service.
Input interface module 202-1 can allocate a number of input bits
equal to the maximum data field capacity of a Baseband (BB) frame.
A padding may be inserted to complete the LDPC/BCH code block
capacity. The input stream sync module 203-1 can provide a
mechanism to regenerate, in the receiver, the clock of the
Transport Stream (or packetized Generic Stream), in order to
guarantee end-to-end constant bit rates and delay.
In order to allow the Transport Stream recombining without
requiring additional memory in the receiver, the input Transport
Streams are delayed by delay compensators 204-1.about.n considering
interleaving parameters of the data PLPs in a group and the
corresponding common PLP. Null packet deleting modules
205-1.about.n can increase transmission efficiency by removing
inserted null packet for a case of VBR (variable bit rate) service.
Cyclic Redundancy Check (CRC) encoder modules 206-1.about.n can add
CRC parity to increase transmission reliability of BB frame. BB
header inserting modules 207-1.about.n can add BB frame header at a
beginning portion of BB frame. Information that can be included in
BB header is shown in FIG. 36.
A Merger/slicer module 208 can perform BB frame slicing from each
PLP, merging BB frames from multiple PLPs, and scheduling each BB
frame within a transmission frame. Therefore, the merger/slicer
module 208 can output L1 signaling information which relates to
allocation of PLP in frame. Lastly, a BB scrambler module 209 can
randomize input bitstreams to minimize correlation between bits
within bitstreams. The modules in shadow in FIG. 35 are modules
used when transmission system uses a single PLP, the other modules
in FIG. 35 are modules used when the transmission device uses
multiple PLPs.
FIG. 37 shows an example of BICM module. FIG. 37a shows data path
and FIG. 37b shows L1 path of BICM module. An outer coder module
301 and an inner coder module 303 can add redundancy to input
bitstreams for error correction. An outer interleaver module 302
and an inner interleaver module 304 can interleave bits to prevent
burst error. The Outer interleaver module 302 can be omitted if the
BICM is specifically for DVB-C2. A bit demux module 305 can control
reliability of each bit output from the inner interleaver module
304. A symbol mapper module 306 can map input bitstreams into
symbol streams. At this time, it is possible to use any of a
conventional QAM, an MQAM which uses the aforementioned BRGC for
performance improvement, an NU-QAM which uses Non-uniform
modulation, or an NU-MQAM which uses Non-uniform modulation applied
BRGC for performance improvement. To construct a system which is
more robust against noise, combinations of modulations using MQAM
and/or NU-MQAM depending on the code rate of the error correction
code and the constellation capacity can be considered. At this
time, the Symbol mapper module 306 can use a proper constellation
according to the code rate and constellation capacity. FIG. 39
shows an example of such combinations.
Case 1 shows an example of using only NU-MQAM at low code rate for
simplified system implementation. Case 2 shows an example of using
optimized constellation at each code rate. The transmitter can send
information about the code rate of the error correction code and
the constellation capacity to the receiver such that the receiver
can use an appropriate constellation. FIG. 40 shows another example
of cases where compatibility between conventional systems is
considered. In addition to the examples, further combinations for
optimizing the system are possible.
The ModCod Header inserting module 307 shown in FIG. 37 can take
Adaptive coding and modulation (ACM)/Variable coding and modulation
(VCM) feedback information and add parameter information used in
coding and modulation to a FEC block as header. The Modulation
type/Coderate (ModCod) header can include the following
information:
*FEC type (1 bits)-long or short LDPC
*Coderate (3 bits)
*Modulation (3 bits)-up-to 64K QAM
*PLP identifier (8 bits)
The Symbol interleaver module 308 can perform interleaving in
symbol domain to obtain additional interleaving effects. Similar
processes performed on data path can be performed on L1 signaling
path but with possibly different parameters (301-1.about.308-1). At
this point, a shortened/punctured code module (303-1) can be used
for inner code.
FIG. 38 shows an example of LDPC encoding using
shortening/puncturing. Shortening process can be performed on input
blocks which have less bits than a required number of bits for LDPC
encoding as many zero bits required for LDPC encoding can be padded
(301c). Zero Padded input bitstreams can have parity bits through
LDPC encoding (302c). At this time, for bitstreams that correspond
to original bitstreams, zeros can be removed (303c) and for parity
bitstreams, puncturing (304c) can be performed according to
code-rates. These processed information bitstreams and parity
bitstreams can be multiplexed into original sequences and outputted
(305c).
FIG. 41 shows a frame structure which comprises preamble for L1
signaling and data symbol for PLP data. It can be seen that
preamble and data symbols are cyclically generated, using one frame
as a unit. Data symbols comprise PLP type 0 which is transmitted
using a fixed modulation/coding and PLP type 1 which is transmitted
using a variable modulation/coding. For PLP type 0, information
such as modulation, FEC type, and FEC code rate are transmitted in
preamble (see FIG. 42 Frame header insert 401). For PLP type 1,
corresponding information can be transmitted in FEC block header of
a data symbol (see FIG. 37 ModCod header insert 307). By the
separation of PLP types, ModCod overhead can be reduced by
3.about.4% from a total transmission rate, for PLP type0 which is
transmitted at a fixed bit rate. At a receiver, for fixed
modulation/coding PLP of PLP type 0, Frame header remover r401
shown in FIG. 63 can extract information on Modulation and FEC code
rate and provide the extracted information to a BICM decoding
module. For variable modulation/coding PLP of PLP type 1, ModCod
extracting modules, r307 and r307-1 shown in FIG. 64 can extract
and provide the parameters necessary for BICM decoding.
FIG. 42 shows an example of a frame builder. A frame header
inserting module 401 can form a frame from input symbol streams and
can add frame header at front of each transmitted frame. The frame
header can include the following information:
*Number of bonded channels (4 bits)
*Guard interval (2 bits)
*PAPR (2 bits)
*Pilot pattern (2 bits)
*Digital System identification (16 bits)
*Frame identification (16 bits)
*Frame length (16 bits) number of Orthogonal Frequency Division
Multiplexing (OFDM) symbols per frame
*Superframe length (16 bits) number of frames per superframe
*number of PLPs (8 bits)
*for each PLP
PLP identification (8 bits)
Channel bonding id (4 bits)
PLP start (9 bits)
PLP type (2 bits) common PLP or others
PLP payload type (5 bits)
MC type (1 bit)-fixed/variable modulation & coding
if MC type=fixed modulation & coding
FEC type (1 bits)-long or short LDPC
Coderate (3 bits)
Modulation (3 bits)-up-to 64K QAM
end if;
Number of notch channels (2 bits)
for each notch
Notch start (9 bits)
Notch width (9 bits)
end for;
PLP width (9 bits)-max number of FEC blocks of PLP
PLP time interleaving type (2 bits)
end for;
*CRC-32 (32 bits)
Channel bonding environment is assumed for L1 information
transmitted in Frame header and data that correspond to each data
slice is defined as PLP. Therefore, information such as PLP
identifier, channel bonding identifier, and PLP start address are
required for each channel used in bonding. One embodiment of this
invention suggests transmitting ModCod field in FEC frame header if
PLP type supports variable modulation/coding and transmitting
ModCod field in Frame header if PLP type supports fixed
modulation/coding to reduce signaling overhead. In addition, if a
Notch band exists for each PLP, by transmitting the start address
of the Notch and its width, decoding corresponding carriers at the
receiver can become unnecessary.
FIG. 43 shows an example of Pilot Pattern 5 (PP5) applied in a
channel bonding environment. As shown, if SP positions are
coincident with preamble pilot positions, irregular pilot structure
can occur.
FIG. 43a shows an example of pilot inserting module 404 as shown in
FIG. 42. As represented in FIG. 43, if a single frequency band (for
example, 8 MHz) is used, the available bandwidth is 7.61 MHz, but
if multiple frequency bands are bonded, guard bands can be removed,
thus, frequency efficiency can increase greatly. FIG. 43b is an
example of preamble inserting module 504 as shown in FIG. 51 that
is transmitted at the front part of the frame and even with channel
bonding, the preamble has repetition rate of 7.61 MHz, which is
bandwidth of L1 block. This is a structure considering the
bandwidth of a tuner which performs initial channel scanning.
Pilot Patterns exist for both Preamble and Data Symbols. For data
symbol, scattered pilot (SP) patterns can be used. Pilot Pattern 5
(PP5) and Pilot Pattern 7 (PP7) of T2 can be good candidates for
frequency-only interpolation. PP5 has x=12, y=4, z=48 for GI= 1/64
and PP7 has x=24, y=4, z=96 for GI= 1/128. Additional
time-interpolation is also possible for a better channel
estimation. Pilot patterns for preamble can cover all possible
pilot positions for initial channel acquisition. In addition,
preamble pilot positions should be coincident with SP positions and
a single pilot pattern for both the preamble and the SP is desired.
Preamble pilots could also be used for time-interpolation and every
preamble could have an identical pilot pattern. These requirements
are important for C2 detection in scanning and necessary for
frequency offset estimation with scrambling sequence correlation.
In a channel bonding environment, the coincidence in pilot
positions should also be kept for channel bonding because irregular
pilot structure may degrade interpolation performance.
In detail, if a distance z between scattered pilots (SPs) in an
OFDM symbol is 48 and if a distance y between SPs corresponding to
a specific SP carrier along the time axis is 4, an effective
distance x after time interpolation becomes 12. This is when a
guard interval (GI) fraction is 1/64. If GI fraction is 1/128,
x=24, y=4, and z=96 can be used. If channel bonding is used, SP
positions can be made coincident with preamble pilot positions by
generating non-continuous points in scattered pilot structure.
At this time, preamble pilot positions can be coincident with every
SP positions of data symbol. When channel bonding is used, data
slice where a service is transmitted, can be determined regardless
of 8 MHz bandwidth granularity. However, for reducing overhead for
data slice addressing, transmission starting from SP position and
ending at SP position can be chosen.
When a receiver receives such SPs, if necessary, channel estimation
module r501 shown in FIG. 62 can perform time interpolation to
obtain pilots shown in dotted lines in FIG. 43 and perform
frequency interpolation. At this time, for non-continuous points of
which intervals are designated as 32 in FIG. 43, either performing
interpolations on left and right separately or performing
interpolations on only one side then performing interpolation on
the other side by using the already interpolated pilot positions of
which interval is 12 as a reference point can be implemented. At
this time, data slice width can vary within 7.61 MHz, thus, a
receiver can minimize power consumption by performing channel
estimation and decoding only necessary subcarriers.
FIG. 44 shows another example of PP5 applied in channel bonding
environment or a structure of SP for maintaining effective distance
x as 12 to avoid irregular SP structure shown in FIG. 43 when
channel bonding is used. FIG. 44a is a structure of SP for data
symbol and FIG. 44b is a structure of SP for preamble symbol.
As shown, if SP distance is kept consistent in case of channel
bonding, there will be no problem in frequency interpolation but
pilot positions between data symbol and preamble may not be
coincident. In other words, this structure does not require
additional channel estimation for an irregular SP structure,
however, SP positions used in channel bonding and preamble pilot
positions become different for each channel.
FIG. 45 shows a new SP structure or PP5 to provide a solution to
the two problems aforementioned in channel bonding environment.
Specifically, a pilot distance of x=16 can solve those problems. To
preserve pilot density or to maintain the same overhead, a PP5' can
have x=16, y=3, z=48 for GI= 1/64 and a PP7' can have x=16, y=6,
z=96 for GI= 1/128. Frequency-only interpolation capability can
still be maintained. Pilot positions are depicted in FIG. 45 for
comparison with PP5 structure.
FIG. 46 shows an example of a new SP Pattern or PP5 structure in
channel bonding environment. As shown in FIG. 46, whether either
single channel or channel bonding is used, an effective pilot
distance x=16 can be provided. In addition, because SP positions
can be made coincident with preamble pilot positions, channel
estimation deterioration caused by SP irregularity or
non-coincident SP positions can be avoided. In other words, no
irregular SP position exists for freq-interpolator and coincidence
between preamble and SP positions is provided.
Consequently, the proposed new SP patterns can be advantageous in
that single SP pattern can be used for both single and bonded
channel; no irregular pilot structure can be caused, thus a good
channel estimation is possible; both preamble and SP pilot
positions can be kept coincident; pilot density can be kept the
same as for PP5 and PP7 respectively; and Frequency-only
interpolation capability can also be preserved.
In addition, the preamble structure can meet the requirements such
as preamble pilot positions should cover all possible SP positions
for initial channel acquisition; maximum number of carriers should
be 3409 (7.61 MHz) for initial scanning; exactly same pilot
patterns and scrambling sequence should be used for C2 detection;
and no detection-specific preamble like P1 in T2 is required.
In terms of relation with frame structure, data slice position
granularity may be modified to 16 carriers rather than 12, thus,
less position addressing overhead can occur and no other problem
regarding data slice condition, Null slot condition etc can be
expected.
Therefore, at channel estimation module r501 of FIG. 62, pilots in
every preamble can be used when time interpolation of SP of data
symbol is performed. Therefore, channel acquisition and channel
estimation at the frame boundaries can be improved.
Now, regarding requirements related to the preamble and the pilot
structure, there is consensus in that positions of preamble pilots
and SPs should coincide regardless of channel bonding; the number
of total carriers in L1 block should be dividable by pilot distance
to avoid irregular structure at band edge; L1 blocks should be
repeated in frequency domain; and L1 blocks should always be
decodable in arbitrary tuner window position. Additional
requirements would be that pilot positions and patterns should be
repeated by period of 8 MHz; correct carrier frequency offset
should be estimated without channel bonding knowledge; and L1
decoding (re-ordering) is impossible before the frequency offset is
compensated.
FIG. 47 shows a relationship between data symbol and preamble when
preamble structures as shown in FIG. 52 and FIG. 53 are used. L1
block can be repeated by period of 6 MHz. For L1 decoding, both
frequency offset and Preamble shift pattern should be found. L1
decoding is not possible in arbitrary tuner position without
channel bonding information and a receiver cannot differentiate
between preamble shift value and frequency offset.
Thus, a receiver, specifically for Frame header remover r401 shown
in FIG. 63 to perform L1 signal decoding, channel bonding structure
needs to be obtained. Because preamble shift amount expected at two
vertically shadowed regions in FIG. 47 is known, time/freq
synchronizing module r505 in FIG. 62 can estimate carrier frequency
offset. Based on the estimation, L1 signaling path
(r308-1.about.r301-1) in FIG. 64 can decode L1.
FIG. 48 shows a relationship between data symbol and preamble when
the preamble structure as shown in FIG. 55 is used. L1 block can be
repeated by period of 8 MHz. For L1 decoding, only frequency offset
needs to be found and channel bonding knowledge may not be
required. Frequency offset can be easily estimated by using known
Pseudo Random Binary Sequence (PRBS) sequence. As shown in FIG. 48,
preamble and data symbols are aligned, thus, additional sync search
can become unnecessary. Therefore, for a receiver, specifically for
the Frame header remover module r401 shown in FIG. 63, it is
possible that only correlation peak with pilot scrambling sequence
needs to be obtained to perform L1 signal decoding. The time/freq
synchronizing module r505 in FIG. 62 can estimate carrier frequency
offset from peak position.
FIG. 49 shows an example of cable channel delay profile.
From the point of view of pilot design, current GI already
over-protects delay spread of cable channel. In the worst case,
redesigning the channel model can be an option. To repeat the
pattern exactly every 8 MHz, the pilot distance should be a divisor
of 3584 carriers (z=32 or 56). A pilot density of z=32 can increase
pilot overhead, thus, z=56 can be chosen. Slightly less delay
coverage may not be an important in cable channel. For example, it
can be 8 .mu.s for PP5' and 4 .mu.s for PP7' compared to 9.3 .mu.s
(PP5) and 4.7 .mu.s (PP7). Meaningful delays can be covered by both
pilot patterns even in a worst case. For preamble pilot position,
no more than all SP positions in data symbol are necessary.
If the -40 dB delay path can be ignored, actual delay spread can
become 2.5 us, 1/64 GI=7 .mu.s, or 1/128 GI=3.5 is. This shows that
pilot distance parameter, z=56 can be a good enough value. In
addition, z=56 can be a convenient value for structuring pilot
pattern that enables preamble structure shown in FIG. 48.
FIG. 50 shows scattered pilot structure that uses z=56 and z=112
which is constructed at pilot inserting module 404 in FIG. 42. PP5'
(x=14, y=4, z=56) and PP7' (x=28, y=4, z=112) are proposed. Edge
carriers could be inserted for closing edge.
As shown in FIG. 50, pilots are aligned at 8 MHz from each edge of
the band, every pilot position and pilot structure can be repeated
every 8 MHz. Thus, this structure can support the preamble
structure shown in FIG. 48. In addition, a common pilot structure
between preamble and data symbols can be used. Therefore, channel
estimation module r501 in FIG. 62 can perform channel estimation
using interpolation on preamble and data symbols because no
irregular pilot pattern can occur, regardless of window position
which is decided by data slice locations. At this time, using only
frequency interpolation can be enough to compensate channel
distortion from delay spread. If time interpolation is performed
additionally, more accurate channel estimation can be
performed.
Consequently, in the new proposed pilot pattern, pilot position and
pattern can be repeated based on a period of 8 MHz. A single pilot
pattern can be used for both preamble and data symbols. L1 decoding
can always be possible without channel bonding knowledge. In
addition, the proposed pilot pattern may not affect commonality
with T2 because the same pilot strategy of scattered pilot pattern
can be used; T2 already uses 8 different pilot patterns; and no
significant receiver complexity can be increased by modified pilot
patterns. For a pilot scrambling sequence, the period of PRBS can
be 2047 (m-sequence); PRBS generation can be reset every 8 MHz. of
which the period is 3584; pilot repetition rate of 56 can be also
co-prime with 2047; and no PAPR issue can be expected.
FIG. 51 shows an example of a modulator based on OFDM. Input symbol
streams can be transformed into time domain by IFFT module 501. If
necessary, peak-to-average power ratio (PAPR) can be reduced at
PAPR reducing module 502. For PAPR methods, Active constellation
extension (ACE) or tone reservation can be used. GI inserting
module 503 can copy a last part of effective OFDM symbol to fill
guard interval in a form of cyclic prefix.
Preamble inserting module 504 can insert preamble at the front of
each transmitted frame such that a receiver can detect digital
signal, frame and acquire time/freq offset acquisition. At this
time, the preamble signal can perform physical layer signaling such
as FFT size (3 bits) and Guard interval size (3 bits). The Preamble
inserting module 504 can be omitted if the modulator is
specifically for DVB-C2.
FIG. 52 shows an example of a preamble structure for channel
bonding, generated at preamble inserting module 504 in FIG. 51. One
complete L1 block should be "always decodable" in any arbitrary
7.61 MHz tuning window position and no loss of L1 signaling
regardless of tuner window position should occur. As shown, L1
blocks can be repeated in frequency domain by period of 6 MHz. Data
symbol can be channel bonded for every 8 MHz. If, for L1 decoding,
a receiver uses a tuner such as the tuner r603 represented in FIG.
61 which uses a bandwidth of 7.61 MHz, the Frame header remover
r401 in FIG. 63 needs to rearrange the received cyclic shifted L1
block (FIG. 53) to its original form. This rearrangement is
possible because L1 block is repeated for every 6 MHz block. FIG.
53a can be reordered into FIG. 53b.
FIG. 54 shows a process for designing a more optimized preamble.
The preamble structure of FIG. 52 uses only 6 MHz of total tuner
bandwidth 7.61 MHz for L1 decoding. In terms of spectrum
efficiency, tuner bandwidth of 7.61 MHz is not fully utilized.
Therefore, there can be further optimization in spectrum
efficiency.
FIG. 55 shows another example of preamble structure or preamble
symbols structure for full spectrum efficiency, generated at Frame
Header Inserting module 401 in FIG. 42. Just like data symbol, L1
blocks can be repeated in frequency domain by period of 8 MHz. One
complete L1 block is still "always decodable" in any arbitrary 7.61
MHz tuning window position. After tuning, the 7.61 MHz data can be
regarded as a virtually punctured code. Having exactly the same
bandwidth for both the preamble and data symbols and exactly the
same pilot structure for both the preamble and data symbols can
maximize spectrum efficiency. Other features such as cyclic shifted
property and not sending L1 block in case of no data slice can be
kept unchanged. In other words, the bandwidth of preamble symbols
can be identical with the bandwidth of data symbols or, as shown in
FIG. 57, the bandwidth of the preamble symbols can be the bandwidth
of the tuner (here, it's 7.61 MHz). The tuner bandwidth can be
defined as a bandwidth that corresponds to a number of total active
carriers when a single channel is used. That is, the bandwidth of
the preamble symbol can correspond to the number of total active
carriers (here, it's 7.61 MHz).
FIG. 56 shows a virtually punctured code. The 7.61 MHz data among
the 8 MHz L1 block can be considered as punctured coded. When a
tuner r603 shown in FIG. 61 uses 7.61 MHz bandwidth for L1
decoding, Frame header remover r401 in FIG. 63 needs to rearrange
received, cyclic shifted L1 block into original form as shown in
FIG. 56. At this time, L1 decoding is performed using the entire
bandwidth of the tuner. Once the L1 block is rearranged, a spectrum
of the rearranged L1 block can have a blank region within the
spectrum as shown in upper right side of FIG. 56 because an
original size of L1 block is 8 MHz bandwidth.
Once the blank region is zero padded, either after deinterleaving
in symbol domain by the freq. deinterleaver r403 in FIG. 63 or by
the symbol deinterleaver r308-1 in FIG. 64 or after deinterleaving
in bit domain by the symbol demapper r306-1, bit mux r305-1, and
inner deinterleaver r304-1 in FIG. 64, the block can have a form
which appears to be punctured as shown in lower right side of FIG.
56.
This L1 block can be decoded at the punctured/shortened decode
module r303-1 in FIG. 64. By using these preamble structure, the
entire tuner bandwidth can be utilized, thus spectrum efficiency
and coding gain can be increased. In addition, an identical
bandwidth and pilot structure can be used for the preamble and data
symbols.
In addition, if the preamble bandwidth or the preamble symbols
bandwidth is set as a tuner bandwidth as shown in FIG. 58, (it's
7.61 MHz in the example), a complete L1 block can be obtained after
rearrangement even without puncturing. In other words, for a frame
having preamble symbols, wherein the preamble symbols have at least
one layer 1 (L1) block, it can be said, the L1 block has 3408
active subcarriers and the 3408 active subcarriers correspond to
7.61 MHz of 8 MHz Radio Frequency (RF) band.
Thus, spectrum efficiency and L1 decoding performance can be
maximized. In other words, at a receiver, decoding can be performed
at punctured/shortened decode module r303-1 in FIG. 64, after
performing only deinterleaving in the symbol domain.
Consequently, the proposed new preamble structure can be
advantageous in that it's fully compatible with previously used
preamble except that the bandwidth is different; L1 blocks are
repeated by period of 8 MHz; L1 block can be always decodable
regardless of tuner window position; Full tuner bandwidth can be
used for L1 decoding; maximum spectrum efficiency can guarantee
more coding gain; incomplete L1 block can be considered as
punctured coded; simple and same pilot structure can be used for
both preamble and data; and identical bandwidth can be used for
both preamble and data.
FIG. 59 shows an example of an analog processor. A DAC module 601
can convert digital signal input into analog signal. After
transmission frequency bandwidth is up-converted (602) and analog
filtered (603) signal can be transmitted.
FIG. 60 shows an example of a digital receiver system. Received
signal is converted into digital signal at an analog process module
r105. A demodulator r104 can convert the signal into data in
frequency domain. A frame parser r103 can remove pilots and headers
and enable selection of service information that needs to be
decoded. A BICM demodulator r102 can correct errors in the
transmission channel. An output processor r101 can restore the
originally transmitted service stream and timing information.
FIG. 61 shows an example of analog processor used at the receiver.
A Tuner/AGC module r603 can select desired frequency bandwidth from
received signal. A down converting module r602 can restore
baseband. An ADC module r601 can convert analog signal into digital
signal.
FIG. 62 shows an example of demodulator. A frame detecting module
r506 can detect the preamble, check if a corresponding digital
signal exists, and detect a start of a frame. A time/freq
synchronizing module r505 can perform synchronization in time and
frequency domains. At this time, for time domain synchronization, a
guard interval correlation can be used. For frequency domain
synchronization, correlation can be used or offset can be estimated
from phase information of a subcarrier that is transmitted in the
frequency domain. A preamble removing module r504 can remove
preamble from the front of detected frame. A GI removing module
r503 can remove guard interval. A FFT module r501 can transform
signal in the time domain into signal in the frequency domain. A
channel estimation/equalization module r501 can compensate errors
by estimating distortion in transmission channel using pilot
symbol. The Preamble removing module r504 can be omitted if the
demodulator is specifically for DVB-C2.
FIG. 63 shows an example of frame parser. A pilot removing module
r404 can remove pilot symbol. A freq deinterleaving module r403 can
perform deinterleaving in the frequency domain. An OFDM symbol
merger r402 can restore data frame from symbol streams transmitted
in OFDM symbols. A frame header removing module r401 can extract
physical layer signaling from header of each transmitted frame and
remove header. Extracted information can be used as parameters for
following processes in the receiver.
FIG. 64 shows an example of a BICM demodxulator. FIG. 64a shows a
data path and FIG. 64b shows a L1 signaling path. A symbol
deinterleaver r308 can perform deinterleaving in the symbol domain.
A ModCod extract r307 can extract ModCod parameters from front of
each BB frame and make the parameters available for following
adaptive/variable demodulation and decoding processes. A Symbol
demapper r306 can demap input symbol streams into bit
Log-Likelyhood Ratio (LLR) streams. The Output bit LLR streams can
be calculated by using a constellation used in a Symbol mapper 306
of the transmitter as reference point. At this point, when the
aforementioned MQAM or NU-MQAM is used, by calculating both I axis
and Q axis when calculating bit nearest from MSB and by calculating
either I axis or Q axis when calculating the rest bits, an
efficient symbol demapper can be implemented. This method can be
applied to, for example, Approximate LLR, Exact LLR, or Hard
decision.
When an optimized constellation according to constellation capacity
and code rate of error correction code at the Symbol mapper 306 of
the transmitter is used, the Symbol demapper r306 of the receiver
can obtain a constellation using the code rate and constellation
capacity information transmitted from the transmitter. The bit mux
r305 of the receiver can perform an inverse function of the bit
demux 305 of the transmitter. The Inner deinterleaver r304 and
outer deinterleaver r302 of the receiver can perform inverse
functions of the inner interleaver 304 and outer interleaver 302 of
the transmitter, respectively to get the bitstream in its original
sequence. The outer deinterleaver r302 can be omitted if the BICM
demodulator is specifically for DVB-C2.
The inner decoder r303 and outer decoder r301 of the receiver can
perform corresponding decoding processes to the inner coder 303 and
outer code 301 of the transmitter, respectively, to correct errors
in the transmission channel. Similar processes performed on data
path can be performed on L1 signaling path, but with different
parameters (r308-1.about.r301-1). At this point, as explained in
the preamble part, a shortened/punctured code module r303-1 can be
used for L1 signal decoding.
FIG. 65 shows an example of IDPC decoding using
shortening/puncturing. A demux r301a can separately output
information part and parity part of systematic code from input bit
streams. For the information part, a zero padding (r302a) can be
performed according to a number of input bit streams of LDPC
decoder, for the parity part, input bit streams for (r303a) the
LDPC decoder can be generated by depuncturing punctured part. LDPC
decoding (r304a) can be performed on generated bit streams, zeros
in information part can be removed and output (r305a).
FIG. 66 shows an example of output processor. A BB descrambler r209
can restore scrambled (209) bit streams at the transmitter. A
Splitter r208 can restore BB frames that correspond to multiple PLP
that are multiplexed and transmitted from the transmitter according
to PLP path. For each PLP path, a BB header remover r207-1.about.n
can remove the header that is transmitted at the front of the BB
frame. A CRC decoder r206-1.about.n can perform CRC decoding and
make reliable BB frames available for selection. A Null packet
inserting modules r205-1.about.n can restore null packets which
were removed for higher transmission efficiency in their original
location. A Delay recovering modules r204-1.about.n can restore a
delay that exists between each PLP path.
An output clock recovering modules r203-1.about.n can restore the
original timing of the service stream from timing information
transmitted from the input stream synchronization modules
203-1.about.n. An output interface modules r202-1.about.n can
restore data in TS/GS packet from input bit streams that are sliced
in BB frame. An output postprocess modules r201-1.about.n can
restore multiple TS/GS streams into a complete TS/GS stream, if
necessary. The shaded blocks shown in FIG. 66 represent modules
that can be used when a single PLP is processed at a time and the
rest of the blocks represent modules that can be used when multiple
PLPs are processed at the same time.
Preamble pilot patterns were carefully designed to avoid PAPR
increase, thus, whether L1 repetition rate may increase PAPR needs
to be considered. The number of L1 information bits varies
dynamically according to the channel bonding, the number of PLPs,
etc. In detail, it is necessary to consider things such as fixed L1
block size may introduce unnecessary overhead; L1 signaling should
be protected more strongly than data symbols; and time interleaving
of L1 block can improve robustness over channel impairment such as
impulsive noise need.
For a L1 block repetition rate of 8 MHz, as shown in FIG. 67, full
spectrum efficiency (26.8% BW increase) is exhibited with virtual
puncturing but the PAPR may be increased since L1 bandwidth is the
same as that of the data symbols. For the repetition rate of 8 MHz,
4K-FFT DVB-T2 frequency interleaving can be used for commonality
and the same pattern can repeat itself at a 8 MHz period after
interleaving.
For a L1 block repetition rate of 6 MHz, as shown in FIG. 68,
reduced spectrum efficiency can be exhibited with no virtual
puncturing. A similar problem of PAPR as for the 8 MHz case can
occur since the L1 and data symbol bandwidths share LCM=24 MHz. For
the repetition rate of 6 MHz, 4K-FFT DVB-T2 frequency interleaving
can be used for commonality and the same pattern can repeat itself
at a period of 24 MHz after interleaving.
FIG. 69 shows a new L1 block repetition rate of 7.61 MHz or full
tuner bandwidth. A full spectrum efficiency (26.8% BW increase) can
be obtained with no virtual puncturing. There can be no PAPR issue
since L1 and data symbol bandwidths share LCM=1704 MHz. For the
repetition rate of 7.61 MHz, 4K-FFT DVB-T2 frequency interleaving
can be used for commonality and the same pattern can repeat itself
by period of about 1704 MHz after interleaving.
FIG. 70 is an example of L1 signaling which is transmitted in the
frame header. Each information in L1 signaling can be transmitted
to the receiver and can be used as a decoding parameter.
Especially, the information can be used in L1 signal path shown in
FIG. 64 and PLPs can be transmitted in each data slice. An
increased robustness for each PLP can be obtained.
FIG. 72 is an example of a symbol interleaver 308-1 as shown in L1
signaling path in FIG. 37 and also can be an example of its
corresponding symbol deinterleaver r308-1 as shown in L1 signaling
path in FIG. 64. Blocks with tilted lines represent L1 blocks and
solid blocks represent data carriers. L1 blocks can be transmitted
not only within a single preamble, but also can be transmitted
within multiple OFDM blocks. Depending on a size of L1 block, the
size of the interleaving block can vary. In other words, num_L1_sym
and L1 span can be different from each other. To minimize
unnecessary overhead, data can be transmitted within the rest of
the carriers of the OFDM symbols where the L1 block is transmitted.
At this point, full spectrum efficiency can be guaranteed because
the repeating cycle of L1 block is still a full tuner bandwidth. In
FIG. 72, the numbers in blocks with tilted lines represent the bit
order within a single LDPC block.
Consequently, when bits are written in an interleaving memory in
the row direction according to a symbol index as shown in FIG. 72
and read in the column direction according to a carrier index, a
block interleaving effect can be obtained. In other words, one LDPC
block can be interleaved in the time domain and the frequency
domain and then can be transmitted. Num_L1_sym can be a
predetermined value, for example, a number between 2.about.4 can be
set as a number of OFDM symbols. At this point, to increase the
granularity of the L1 block size, a punctured/shortened LDPC code
having a minimum length of the codeword can be used for L1
protection.
FIG. 73 is an example of an L1 block transmission. FIG. 73
illustrates FIG. 72 in frame domain. As shown on FIG. 73a, L1
blocks can be spanning in full tuner bandwidth or as shown on FIG.
73b, L1 blocks can be partially spanned and the rest of the
carriers can be used for data carrier. In either case, it can be
seen that the repetition rate of L1 block can be identical to a
full tuner bandwidth. In addition, for OFDM symbols which uses L1
signaling including preamble, only symbol interleaving can be
performed while not allowing data transmission in that OFDM
symbols. Consequently, for OFDM symbol used for L1 signaling, a
receiver can decode L1 by performing deinterleaving without data
decoding. At this point, the L1 block can transmit L1 signaling of
current frame or L1 signaling of a subsequent frame. At the
receiver side, L1 parameters decoded from L1 signaling decoding
path shown in FIG. 64 can be used for decoding process for data
path from frame parser of subsequent frame.
In summary, at a transmitter, interleaving blocks of L1 region can
be performed by writing blocks to a memory in a row direction and
reading the written blocks from the memory in a column direction.
At a receiver, deinterleaving blocks of L1 region can be performed
by writing blocks to a memory in a column direction and reading the
written blocks from the memory in a row direction. The reading and
writing directions of transmitter and receiver can be
interchanged.
When simulation is performed with assumptions such as CR=1/2 for L1
protection and for T2 commonality; 16-QAM symbol mapping; pilot
density of 6 in the Preamble; number of short LDPC implies required
amount of puncturing/shortening are made, results or conclusions
such as only preamble for L1 transmission may not be sufficient;
the number of OFDM symbols depends on the amount of L1 block size;
shortest LDPC codeword (e.g. 192 bits information) among
shortened/punctured code may be used for flexibility and fine
granularity; and Padding may be added if required with negligible
overhead, can be obtained. The result is summarized in FIG. 71.
Consequently, for a L1 block repetition rate, full tuner bandwidth
with no virtual puncturing can be a good solution and still no PAPR
issue can arise with full spectrum efficiency. For L1 signaling,
efficient signaling structure can allow maximum configuration in an
environment of 8 channels bonding, 32 notches, 256 data slices, and
256 PLPs. For L1 block structure, flexible L1 signaling can be
implemented according to L1 block size. Time interleaving can be
performed for better robustness for T2 commonality. Less overhead
can allow data transmission in preamble.
Block interleaving of L1 block can be performed for better
robustness. The interleaving can be performed with fixed
pre-defined number of L1 symbols (num_L1_sym) and a number of
carriers spanned by L1 as a parameter (L1_span). The same technique
is used for P2 preamble interleaving in DVB-T2.
L1 block of variable size can be used. Size can be adaptable to the
amount of L1 signaling bits, resulting in a reduced overhead. Full
spectrum efficiency can be obtained with no PAPR issue. Less than
7.61 MHz repetition can mean that more redundancy can be sent but
unused. No PAPR issue can arise because of 7.61 MHz repetition rate
for L1 block.
Using the suggested methods and devices, among others advantages it
is possible to implement an efficient digital transmitter, receiver
and structure of physical layer signaling.
By transmitting ModCod information in each BB frame header that is
necessary for ACM/VCM and transmitting the rest of the physical
layer signaling in a frame header, signaling overhead can be
minimized.
Modified QAM for a more energy efficient transmission or a more
noise-robust digital broadcasting system can be implemented. The
system can include transmitter and receiver for each example
disclosed and the combinations thereof.
An Improved Non-uniform QAM for a more energy efficient
transmission or a more noise-robust digital broadcasting system can
be implemented. A method of using code rate of error correction
code of NU-MQAM and MQAM is also described. The system can include
transmitter and receiver for each example disclosed and the
combinations thereof.
The suggested L1 signaling method can reduce overhead by 3.about.4%
by minimizing signaling overhead during channel bonding.
It will be apparent to those skilled in the art that various
modifications and variations can be made in the present invention
without departing from the invention.
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