U.S. patent application number 09/773801 was filed with the patent office on 2001-09-06 for high voltage stabilizing circuit.
Invention is credited to Yasumura, Masayuki.
Application Number | 20010019491 09/773801 |
Document ID | / |
Family ID | 18557234 |
Filed Date | 2001-09-06 |
United States Patent
Application |
20010019491 |
Kind Code |
A1 |
Yasumura, Masayuki |
September 6, 2001 |
High voltage stabilizing circuit
Abstract
A high voltage stabilizing circuit according to the present
invention has a voltage resonance type converter as a switching
converter that performs switching operation using direct-current
input voltage obtained from commercial alternating-current power,
and switching output of the voltage resonance type converter is
directly transmitted to a primary winding of a flyback transformer
FBT (high voltage output transformer). Then, a rectifier circuit
for high voltage connected on the secondary side of the flyback
transformer FBT provides high direct-current voltage as an anode
voltage for a CRT, for example. The high direct-current voltage is
stabilized by controlling the switching frequency of the voltage
resonance type converter on the primary side according to the level
of the high direct-current voltage. This configuration requires
only one switching converter to be provided on the primary side of
the flyback transformer FBT.
Inventors: |
Yasumura, Masayuki;
(Kanagawa, JP) |
Correspondence
Address: |
FROMMER LAWRENCE & HAUG LLP
745 FIFTH AVENUE
NEW YORK
NY
10151
US
|
Family ID: |
18557234 |
Appl. No.: |
09/773801 |
Filed: |
February 1, 2001 |
Current U.S.
Class: |
363/21.12 |
Current CPC
Class: |
H02M 1/0058 20210501;
H02M 3/3385 20130101; H02M 1/007 20210501; Y02B 70/10 20130101 |
Class at
Publication: |
363/21.12 |
International
Class: |
H02M 003/335 |
Foreign Application Data
Date |
Code |
Application Number |
Feb 3, 2000 |
JP |
2000-032664 |
Claims
What is claimed is:
1. A high voltage stabilizing circuit comprising: rectifying and
smoothing means for rectifying and smoothing commercial
alternating-current power and thereby providing direct-current
input voltage; switching means having a switching device for
interrupting said direct-current input voltage for output; a
parallel resonant circuit on a primary side formed so as to convert
operation of said switching means into voltage resonance type
operation; a high voltage output transformer having a primary-side
winding to which switching output of said switching means is
transmitted and a secondary-side winding closely coupled to the
primary winding for inducing a required high level of stepped-up
alternating voltage; high direct-current voltage generating means
supplied with said stepped-up alternating voltage for generating
and outputting a required high level of direct-current voltage; and
constant voltage control means for controlling said high
direct-current voltage to a constant voltage by changing switching
frequency of said switching device according to level of said high
direct-current voltage.
2. A high voltage stabilizing circuit as claimed in claim 1,
wherein: said parallel resonant circuit is formed by a choke coil
inserted between a positive terminal of a smoothing capacitor
forming said rectifying and smoothing means and a switching output
point of the switching device, and a parallel resonant capacitor
and a diode device connected in parallel with said switching
device; and a series connection circuit of said primary-side
winding and a direct current blocking capacitor is connected to
said switching device.
3. A high voltage stabilizing circuit as claimed in claim 1,
wherein: said parallel resonant circuit is formed by said
primary-side winding inserted between a positive terminal of a
smoothing capacitor forming said rectifying and smoothing means and
a switching output point of the switching device, and a parallel
resonant capacitor connected in parallel with said switching
device.
Description
BACKGROUND OF THE INVENTION
[0001] The present invention relates to a high voltage stabilizing
circuit which generates stabilized high voltage from commercial
alternating-current power to provide anode voltage to be applied to
an anode electrode of a cathode-ray tube, for example.
[0002] Television receivers, projectors, and display units for
personal computers which employ cathode-ray tubes (hereinafter
referred to as CRTs) as display devices have widely spread.
[0003] As is well known in the art, in various display units having
such CRTs, a required level of high voltage (anode voltage) needs
to be supplied in a stable manner to anode electrodes of the
CRTs.
[0004] Specifically, although the level of anode voltage is
generally set at about 25 KV to 35 KV, the anode voltage is varied
according to variations in alternating-current input voltage and a
load, for example. Variations in the anode voltage may in turn vary
screen sizes in a vertical and a horizontal direction of an image
displayed on the CRT. In addition, variations in the anode voltage
cause a bright white peak image to be distorted when displayed on
the screen. As shown in FIG. 9 for example, when an original
rectangular white peak image as indicated by a solid line is
displayed, the white peak image is distorted into a trapezoidal
shape as indicated by a broken line.
[0005] Therefore, in practice, a so-called high voltage stabilizing
circuit is provided as an anode voltage supplying circuit to
stabilize anode voltage for output.
[0006] FIG. 6 is a circuit diagram showing an example of a high
voltage stabilizing circuit. The high voltage stabilizing circuit
shown in the figure is to be provided for a so-called
multiscanning-capable display unit for a personal computer.
[0007] First, FIG. 6 shows a rectifying and smoothing circuit
comprising a bridge rectifier circuit Di and a smoothing capacitor
Ci. The rectifying and smoothing circuit rectifies and smoothes
commercial alternating-current power AC to provide a rectified and
smoothed voltage Ei whose level is equal to that of the commercial
alternating-current power AC. The rectified and smoothed voltage Ei
is supplied as direct-current input voltage to a switching power
supply circuit 10.
[0008] The switching power supply circuit 10 is a DC-to-DC
converter configured to provide a direct-current output voltage Eo
by switching and stabilizing the inputted rectified and smoothed
voltage Ei. In this case, the switching power supply circuit 10
outputs a direct-current output voltage Eo stabilized at 240 V.
[0009] The direct-current output voltage Eo is inputted to a
step-down converter 20.
[0010] The step-down converter 20 is for example formed by
connecting a drain and a source of a MOS-FET switching device Q11
in series with a choke coil CH1 between a line of the
direct-current output voltage Eo and a positive electrode of a
smoothing capacitor COA and inserting a diode DD1 between a point
where the source of the switching device Q11 and the choke coil CH1
are connected and a primary-side ground.
[0011] The switching device Q11 is externally driven by a driving
voltage from a driving circuit 14, which will be described later,
to perform switching on the direct-current output voltage Eo. A
current that flows according to switching operation by the
switching device Q11 is stored in the smoothing capacitor COA via
the choke coil CH1 and the diode DD1. Then, the step-down converter
20 outputs a stepped-down direct-current voltage EOA, which is a
voltage across the smoothing capacitor COA.
[0012] The stepped-down direct-current voltage EOA is supplied to a
voltage resonance type converter 30.
[0013] The voltage resonance type converter 30 shown in FIG. 6 is
provided with a MOS-FET switching device Q12 to perform externally
excited single-ended operation.
[0014] In the voltage resonance type converter 30, a drain of the
switching device Q12 is connected to a positive terminal of the
stepped-down direct-current voltage EOA via a choke coil CH2, while
a source of the switching device Q12 is connected to the
primary-side ground. A gate of the switching device Q12 is supplied
with a driving voltage outputted from a driving circuit 16, which
will be described later. The switching device Q12 is driven for
switching operation by this driving voltage.
[0015] The drain of the switching device Q12 is also connected to a
starting point of a primary winding N1 of a flyback transformer
FBT, which will be described later. In this case, an ending point
of the primary winding N1 is grounded to the primary-side ground
via a direct current blocking capacitor C11.
[0016] Thus, switching output of the switching device Q12 is
transmitted to the primary winding N1 of the flyback transformer
FBT, whereby an alternating voltage in accordance with the
switching frequency is obtained at the primary winding N1.
[0017] A parallel resonant capacitor Cr is connected in parallel
with the drain and source of the switching device Q12. Capacitance
of the parallel resonant capacitor Cr, inductance L12 of the choke
coil CH2, and leakage inductance L1 of the primary winding N1 side
of the flyback transformer FBT form a primary-side parallel
resonant circuit of the voltage resonance type converter. Although
detailed description is omitted here, voltage V2 across the
resonant capacitor Cr actually forms a sinusoidal pulse waveform
during an off period of the switching device Q12 as a result of the
action of the parallel resonant circuit, so that voltage resonance
type operation is obtained.
[0018] Also, a clamp diode DD2 is connected in parallel with the
drain and source of the switching device Q12. The clamp diode DD2
forms a path of clamp current that flows during the off period of
the switching device Q12.
[0019] The switching devices Q11 and Q12 on the primary side are
driven for switching operation by the following configuration.
[0020] On the basis of a horizontal synchronizing signal frequency
fH corresponding to a currently set resolution, a synchronizing
circuit 11 generates and outputs a horizontal synchronizing signal
having the frequency fH. Since the high voltage stabilizing circuit
in this case is to be provided for a multiscanning display unit,
the horizontal synchronizing signal frequency fH is varied within a
range of 30 KHz to 120 KHz, for example.
[0021] In this case, the horizontal synchronizing signal generated
by the synchronizing circuit 11 is inputted to an oscillating
circuit 12. The oscillating circuit 12 converts the horizontal
synchronizing signal into an oscillating frequency signal to be
used for driving the switching devices Q11 and Q12, and outputs the
converted signal to a PWM control circuit 13 and a driving circuit
16.
[0022] The driving circuit 16 generates a driving voltage for
driving the switching device Q12 from the inputted oscillating
frequency signal, and outputs the driving voltage to the gate of
the switching device Q12. Thus, the switching frequency of the
switching device Q12 coincides with the currently set horizontal
synchronizing signal frequency fH.
[0023] The PWM control circuit 13 effects PWM control of the
inputted oscillating frequency signal on the basis of a detection
output from an error amplifier circuit 15. Specifically, the PWM
control circuit 13 variably controls a duty ratio of an on/off
period in one cycle of the oscillating frequency signal, and
outputs the result to a driving circuit 14. The driving circuit 14
generates driving voltage using a PWMed oscillating frequency
signal outputted from the PWM control circuit 13, and outputs the
signal to the switching device Q11. Thus, the switching frequency
of the switching device Q11 also coincides with the currently set
horizontal synchronizing signal frequency fH. This means that both
the switching device Q11 and the switching device Q12 are allowed
to perform switching operation at a switching frequency in
synchronism with the horizontal synchronizing signal frequency fH.
It should be noted that a duty ratio of an on/off period in one
switching cycle of the switching device Q12 is governed by the
waveform (duty ratio in one cycle) of the oscillating frequency
signal changed by the PWM control circuit 13. This is related to
stabilization of the level of high direct-current voltage EHV, that
is, control for constant voltage, which will be described
later.
[0024] The switching output obtained at the switching device Q12 of
the voltage resonance type converter 30 described above is supplied
to a high voltage generating circuit 40.
[0025] The high voltage generating circuit 40 comprises the flyback
transformer FBT and a high voltage rectifier circuit provided on a
secondary side thereof. The switching output of the switching
device Q12 is transmitted to the primary winding N1 of the flyback
transformer FBT.
[0026] As shown in FIG. 6, the primary winding N1 is wound on the
primary side of the flyback transformer FBT. Also, five step-up
windings NHV1, NHV2, NHV3, NHV4, and NHV5 are wound as secondary
windings on the secondary side of the flyback transformer FBT. In
practice, each of these step-up windings NHV1 to NHV5 is wound
around a core in a divided and independent state. The step-up
windings NHV1 to NHV5 are wound in such a manner that their
polarity is opposite to that of the primary winding N1, whereby
flyback operation is obtained.
[0027] As shown in FIG. 6, the step-up windings NHV1, NHV2, NHV3,
NHV4, and NHV5 are connected in series with high voltage rectifier
diodes DHV1, DHV2, DHV3, DHV4, and DHV5, respectively, thereby
forming a total of five half-wave rectifier circuits. The five
half-wave rectifier circuits are connected in series with each
other. Then, a smoothing capacitor COHV is connected in parallel
with the five serially connected half-wave rectifier circuits.
[0028] Thus, on the secondary side of the flyback transformer FBT,
the five half-wave rectifier circuits rectify voltages induced in
the step-up windings NHV1 to NHV5 to store the rectified voltages
in the smoothing capacitor COHV. Accordingly, the level of
direct-current voltage across the smoothing capacitor COHV is five
times as high as the voltage induced in each of the step-up
windings NHV1 to NHV5. The direct-current voltage across the
smoothing capacitor COHV is converted into high direct-current
voltage EHV to be used as anode voltage for a CRT, for example.
[0029] Operation for constant voltage by the circuit shown in FIG.
6 will next be described.
[0030] A series connection circuit of a voltage dividing resistance
R1-R2 is connected in parallel with the smoothing capacitor COHV at
both terminals thereof provided on the secondary side of the
flyback transformer FBT. Hence, a point of connection between the
voltage dividing resistances R1 and R2 has a voltage level obtained
by dividing high direct-current voltage EHV by their voltage
dividing ratio. The point of connection between the voltage
dividing resistances R1 and R2 is connected to an input of the
error amplifier 15. The error amplifier 15 compares the level of
the divided high direct-current voltage EHV with a predetermined
reference level to detect an error. Specifically, the error
amplifier circuit 15 detects an error of the high direct-current
voltage EHV level with respect to the predetermined reference
level. Then, the error amplifier circuit 15 outputs a direct
current or a direct-current voltage whose level is changed
according to the amount of the error, for example.
[0031] The detection output of the error amplifier 15 is supplied
to the PWM control circuit 13. The PWM control circuit 13 effects
PWM control of the inputted oscillating frequency signal on the
basis of the detection output from the error amplifier 15, and
outputs the result to the driving circuit 14.
[0032] Thus, the switching device Q11 driven by the driving circuit
14 is fixed at the switching frequency in synchronism with the
horizontal synchronizing signal frequency, and performs switching
operation in accordance with the on/off-period duty ratio that is
changed by the PWM control according to variations in the level of
the high direct-current voltage EHV.
[0033] The level of stepped-down direct-current voltage EOA is
determined by the level of the direct-current output voltage Eo,
the switching frequency of the switching device Q11, and the duty
ratio of an on/off period in one switching cycle of the switching
device Q11. The stepped-down direct-current voltage EOA is
expressed as:
EOA=Eo .multidot.TON1/Ts
[0034] where one cycle of switching operation by the switching
device Q11 is Ts and an on period in one cycle is TON1.
[0035] When the duty ratio of an on/off period in one switching
cycle of the switching device Q11 is changed by the PWM control
according to an error of the high direct-current voltage EHV level,
as described above, the level of the stepped-down direct-current
voltage EOA can be variably controlled.
[0036] FIGS. 7A to 7H show operating waveforms of the switching
devices Q11 and Q12 in the circuit shown in FIG. 6. FIGS. 7A to 7D
show operations at a horizontal synchronizing signal frequency
fH=30 KHz, while FIGS. 7E to 7H show operations at the same points
as in FIGS. 7A to 7D respectively at a horizontal synchronizing
signal frequency fH=120 KHz.
[0037] FIGS. 7A and 7E show drain-to-source voltage V1 of the
switching device Q11, and FIGS. 7B and 7F show switching current 11
that flows through the drain of the switching device Q11.
[0038] FIGS. 7C and 7G show voltage V2 across the parallel resonant
capacitor Cr (parallel resonance voltage) connected in parallel
with the switching device Q12, and FIGS. 7D and 7H show switching
current I2 that flows through the drain of the switching device
Q11.
[0039] FIGS. 7A and 7B show operations under PWM control at a
horizontal synchronizing signal frequency fH=30 KHz where one
switching cycle Ts of the switching device Q11 is 3.3 .mu.s, and
the period TON1 during which the switching device Q11 is turned on
is 6.9 .mu.s. In this case, the level of the stepped-down
direct-current voltage EOA is about 50 V.
[0040] FIGS. 7E and 7F show operations under PWM control at a
horizontal synchronizing signal frequency fH=120 KHz where one
switching cycle Ts of the switching device Q11 is 8.3 .mu.s, and
the period TON1 during which the switching device Q11 is turned on
is 0.4 .mu.s. In this case, the level of the stepped-down
direct-current voltage EOA is about 220 V.
[0041] As indicated by operations of the switching device Q12 shown
in FIGS. 7C and 7D and FIGS. 7G and 7H, in the voltage resonance
type converter 30 powered for operation by the stepped-down
direct-current voltage EOA, a period TOFF2 during which the
switching device Q12 is turned off is set to be constant at 3 .mu.s
irrespective of change in the horizontal synchronizing signal
frequency fH. The period TOFF2 can be set to be constant by
selecting the capacitance of the parallel resonant capacitor
Cr.
[0042] When effecting PWM control of switching operation by the
switching device Q11 according to variation in the level of the
high direct-current voltage EHV and setting the off period TOFF2 of
the switching device Q12 to be constant at 3 .mu.s as described
above, it is possible to control the parallel resonance voltage V2
to a constant level irrespective of changes in the horizontal
synchronizing signal frequency fH, as indicated in FIG. 7C.
[0043] The parallel resonance voltage V2 is transmitted as
switching output to the primary winding N1 of the flyback
transformer FBT. Therefore, when the parallel resonance voltage V2
is controlled to a constant level, an alternating voltage obtained
at the primary winding N1 of the flyback transformer FBT is also
maintained at a constant level. Accordingly, the high
direct-current voltage EHV generated on the secondary side of the
flyback transformer FBT is also controlled to a constant level.
[0044] The high voltage stabilizing circuit shown in FIG. 6 thus
stabilizes the level of the high direct-current voltage EHV.
[0045] As a summary of the operations described above,
characteristics of variations of the high direct-current voltage
EHV, the direct-current output voltage Eo, and the stepped-down
direct-current voltage EOA with respect to the horizontal
synchronizing signal frequency fH will be shown in FIG. 8.
[0046] According to the figure, the direct-current output voltage
Eo is stabilized at 240 V by stabilizing operation of the switching
power supply circuit 10.
[0047] The stepped-down direct-current voltage EOA is controlled so
as to change linearly in a range of about 50 V to 220 V for
horizontal synchronizing signal frequencies fH=30 KHz to 120 KHz.
In FIG. 8, stepped-down direct-current voltages EOA when a load
current IHV supplied by the high direct-current voltage EHV to a
load is 1 mA and 0 mA are shown. Although the stepped-down
direct-current voltage EOA at a load current IHV=1 mA is somewhat
lower than the stepped-down direct-current voltage EOA at a load
current IHV=0 mA, substantially the same voltage value is
obtained.
[0048] Thus, by controlling the level of the stepped-down
direct-current voltage EOA as described above and by setting the
parallel resonance voltage V2 generated by the switching operation
of the switching device Q12 to be constant irrespective of change
(switching) of the horizontal synchronizing signal frequency fH as
described above, the high direct-current voltage EHV becomes
constant at about 27 KV, for example.
[0049] The high voltage stabilizing circuit shown in FIG. 6 thus
stabilizes the high direct-current voltage EHV to suppress
variations in screen sizes in a vertical and a horizontal direction
of an image displayed on the CRT.
[0050] In the case of the high voltage stabilizing circuit
configured as shown in FIG. 6, the trapezoidal distortion of the
bright rectangular white image, which is caused by a variation AEHV
in the high direct-current voltage EHV due to an increase in high
voltage load current IHV occurring at the white peak, is dealt with
by increasing the capacitances of the smoothing capacitors CO, COA,
and COHV to a required level.
[0051] However, the high voltage stabilizing circuit configured as
shown in FIG. 6 has the following problems.
[0052] First, the high voltage stabilizing circuit configured as
shown in FIG. 6 performs power conversion in two stages by means of
the step-down converter 20 and the voltage resonance type converter
30 in the process from the input of commercial alternating-current
power to the generation of the high direct-current voltage EHV.
Thus, overall power conversion efficiency in the high voltage
stabilizing circuit as a whole is low; for example, the power
conversion efficiency at a high voltage load power PHV=27 W (=high
direct-current voltage EHV.times.IHV) is actually no more than 72%,
and reactive power in this case is about 10.5 W. Then input power
becomes correspondingly greater.
[0053] Moreover, with such a configuration, the actually assembled
circuit also has a complex structure, thus resulting in increase in
the number of parts.
[0054] Specifically, for example, the circuit requires two ferrite
transformers (an FBT and an insulating converter transformer in the
switching power supply circuit 10), two ferrite choke coils (CH1
and CH2), and at least three switching devices (Q11, Q12, and a
switching device in the switching power supply circuit 10).
Therefore, the size of the circuit itself becomes large as a result
of large mounting area of its printed board, for example, and also
the cost of the circuit becomes correspondingly high. In addition,
the size and cost of parts in the circuit are increased because the
capacitances of the smoothing capacitors CO, COA, and COHV are
increased to suppress the trapezoidal distortion of the bright
rectangular white image, as described above.
SUMMARY OF THE INVENTION
[0055] In view of the problems described above, an object of the
present invention is to provide a high voltage stabilizing circuit
which improves power conversion efficiency and reduces the size and
cost thereof.
[0056] To achieve the above object, according to an aspect of the
present invention, there is provided a high voltage stabilizing
circuit including rectifying and smoothing means for rectifying and
smoothing commercial alternating-current power and thereby
providing direct-current input voltage; switching means having a
switching device for interrupting the direct-current input voltage
for output; a parallel resonant circuit on a primary side formed so
as to convert operation of the switching means into voltage
resonance type operation; a high voltage output transformer having
a primary-side winding to which switching output of the switching
means is transmitted and a secondary-side winding closely coupled
to the primary winding for inducing a required high level of
stepped-up alternating voltage; high direct-current voltage
generating means supplied with the stepped-up alternating voltage
for generating and outputting a required high level of
direct-current voltage; and constant voltage control means for
controlling the high direct-current voltage to a constant voltage
by changing switching frequency of the switching device according
to level of the high direct-current voltage.
[0057] As described above, the high voltage stabilizing circuit
according to the present invention directly transmits switching
output of a voltage resonance type converter provided on the
primary side to the primary side of the high voltage output
transformer, or for example a flyback transformer. A high
direct-current voltage is obtained on the secondary side of the
high voltage output transformer by using the stepped-up alternating
voltage induced by the switching output on the primary side. Also,
the high direct-current voltage is stabilized actively by variably
controlling the switching frequency of the primary-side voltage
resonance type converter according to the level of the high
direct-current voltage.
[0058] Such a configuration requires only one voltage resonance
type converter to serve as a switching converter for power
conversion of direct-current input voltage obtained on the primary
side from commercial alternating-current power.
[0059] The above and other objects, features and advantages of the
present invention will become apparent from the following
description and the appended claims, taken in conjunction with the
accompanying drawings in which like parts or elements denoted by
like reference symbols.
BRIEF DESCRIPTION OF THE DRAWINGS
[0060] FIG. 1 is a circuit configuration diagram of a high voltage
stabilizing circuit according to an embodiment of the present
invention;
[0061] FIGS. 2A to 2D are waveform diagrams showing operation of
main parts in the high voltage stabilizing circuit of the above
embodiment;
[0062] FIG. 3 is a circuit configuration diagram of a high voltage
stabilizing circuit according to another embodiment of the present
invention;
[0063] FIG. 4 is a sectional configuration view of a flyback
transformer provided in the high voltage stabilizing circuit of the
above embodiment, in which step-up windings are wound by a layer
winding method;
[0064] FIG. 5 is a sectional configuration view of a flyback
transformer provided in the high voltage stabilizing circuit of the
above embodiment, in which step-up windings are wound by a division
winding method;
[0065] FIG. 6 is a circuit configuration diagram of a conventional
high voltage stabilizing circuit;
[0066] FIGS. 7A to 7H are waveform diagrams showing operation of
main parts in the high voltage stabilizing circuit of FIG. 6;
[0067] FIG. 8 is a diagram of assistance in explaining levels of
high direct-current voltage and secondary-side direct-current
output voltage with respect to horizontal synchronizing signal
frequency in the high voltage stabilizing circuit of FIG. 6;
and
[0068] FIG. 9 is a diagram of assistance in explaining how a white
peak image displayed on a CRT is distorted.
PREFERRED EMBODIMENTS OF THE INVENTION
[0069] FIG. 1 is a circuit configuration diagram of a high voltage
stabilizing circuit according to an embodiment of the present
invention. In the figure, the same parts as in FIG. 6 are
identified by the same reference numerals, and their description
will be omitted.
[0070] A voltage resonance type converter that includes a switching
device Q1 and performs switching operation by a so-called
single-ended system is provided on a primary side of the circuit
shown in FIG. 1. The voltage resonance type converter serves as a
switching converter that interrupts direct-current input voltage
obtained by rectifying and smoothing commercial alternating-current
power (rectified and smoothed voltage Ei).
[0071] The voltage resonance type converter in this case is
externally excited, and a MOS-FET, for example, is used as the
switching device Q1. A drain of the switching device Q1 is
connected to a positive electrode of a smoothing capacitor Ci via a
choke coil CH, while a source of the switching device Q1 is
connected to a primary-side ground.
[0072] A parallel resonant capacitor Cr is connected in parallel
with the drain and source of the switching device Q1. Also, a clamp
diode (body diode) DD is connected in parallel with the drain and
source of the switching device Q1.
[0073] In this case, capacitance of the parallel resonant capacitor
Cr and inductance L2 of the choke coil CH form a parallel resonant
circuit. The parallel resonant circuit is set to resonate in
response to switching operation by the switching device Q1. Thus,
the switching operation by the switching device Q1 is of a voltage
resonance type.
[0074] The switching device Q1 is driven for switching operation by
an oscillating circuit 4 and a driving circuit 5.
[0075] The oscillating circuit 4 is oscillated to generate and
output an oscillating signal. The driving circuit 5 converts the
oscillating signal into a driving voltage, and outputs the driving
voltage to a gate of the switching device Q1. Thus, the switching
device Q1 is allowed to perform switching operation according to
the oscillating signal generated by the oscillating circuit 4.
Therefore, the switching frequency of the switching device Q1 and a
duty ratio of an on/off period in one switching cycle depend on the
oscillating signal generated by the oscillating circuit 4.
[0076] The oscillating circuit 4 changes the oscillation frequency
of the oscillating signal to be outputted according to an error
detection output supplied from a control circuit 1 via a
photocoupler 2. This means that the switching frequency of the
switching device Q1 is variably controlled, whereby direct-current
high voltage (anode voltage) obtained on a secondary side of a
flyback transformer FBT can be stabilized, as will be described
later.
[0077] In this case, a starting circuit 3 is supplied with the
rectified and smoothed voltage Ei via a starting resistance Rs, and
also supplied with a low direct-current voltage obtained in a
half-wave rectifier circuit comprising a winding N4 provided around
the choke coil CH and a diode D4. The starting circuit 3 is
intended to start the oscillating circuit 4 in response to these
input voltages supplied at the start of power supply, for
example.
[0078] As shown in FIG. 1, the present embodiment has a series
connection circuit [N1-C11] in which a primary winding N1 of the
flyback transformer FBT and a direct current blocking capacitor C11
are connected in series with each other. The series connection
circuit [N1-C11] is connected in parallel with the switching device
Q1. In this case, the drain of the switching device Q1 is connected
to a starting point of the primary winding N1, while the source of
the switching device Q1 is connected to an ending point of the
primary winding N1 via the direct current blocking capacitor
C1.
[0079] Thus, in the present embodiment, switching output of the
switching device Q1 in the primary-side voltage resonance type
converter is directly transmitted to the primary winding N1 of the
flyback transformer FBT.
[0080] Examples of the structure of the flyback transformer FBT in
the present embodiment are shown in section in FIGS. 4 and 5.
[0081] In a first example of the flyback transformer FBT shown in
FIG. 4, two U-shaped cores CR1 and CR2 are combined in such a
manner that magnetic legs of the two U-shaped cores are opposed to
each other, thereby forming a U-U-shaped core CR3. Also, gaps G1
and G2 are provided in portions where ends of the magnetic legs of
the U-shaped core CR1 are opposed to ends of the magnetic legs of
the U-shaped core CR2.
[0082] As shown in FIG. 4, a low voltage winding bobbin LB and a
high voltage winding bobbin HB are attached to one magnetic leg of
the U-U-shaped core CR3. A primary winding N1 and step-up windings
NHV (1 to 5) are separately wound around the low voltage winding
bobbin LB and the high voltage winding bobbin HB, respectively.
[0083] In this case, the primary winding N1 is wound around the low
voltage winding bobbin LB, and the step-up windings NHV are wound
around the high voltage winding bobbin HB. When the step-up
windings NHV (1 to 5) are wound around the high voltage winding
bobbin HB, the step-up windings NHV need to be insulated from each
other. Therefore, the step-up windings NHV are wound by a method
referred to as layer winding, in which an interlayer film F is
inserted between every two of the winding layers formed by the
step-up windings NHV having given numbers of turns.
[0084] In order to insulate the step-up windings NHV (1 to 5) from
each other, a so-called division winding (slit winding) structure
as shown in FIG. 5 may be adopted in addition to the structure
shown in FIG. 4. Incidentally, in FIG. 5, the same parts as in FIG.
4 are identified by the same reference numerals, and their
description will be omitted.
[0085] When the step-up windings NHV are wound by the division
winding method, dividers DV are formed inside a high voltage
winding bobbin HB1 integrally therewith, as shown in FIG. 5. Thus,
a plurality of slits S, or winding regions, are formed between
adjacent dividers DV. Then, the step-up windings NHV are wound in
their respective slits S, whereby the step-up windings NHV are
insulated from each other.
[0086] Incidentally, each of FIGS. 4 and 5 shows a tertiary winding
N3 provided in a portion of a divided region of the low voltage
winding bobbin LB, because in some cases, for example, provision of
low direct-current voltage on the primary side requires the
tertiary winding N3 that generates low alternating voltage, that
is, a source of the low direct-current voltage.
[0087] In the structures of the flyback transformers FBT shown in
FIGS. 4 and 5 adopted in the present embodiment, the primary
winding N1 on the primary side and the step-up windings NHV (1 to
5) on the secondary side are closely coupled to each other.
[0088] Description will hereinafter be made with reference to FIG.
1 again.
[0089] As described above, the switching output of the switching
device Q1 is transmitted to the primary winding N1 of the flyback
transformer FBT. A high voltage generating circuit 40 including the
flyback transformer FBT uses alternating voltage of the primary
winding N1 generated by the transmitted switching output to
ultimately generate and output high direct-current voltage EHV. In
the present embodiment and other examples, the high direct-current
voltage EHV is used as anode voltage of a CRT.
[0090] The circuit configurations of the flyback transformer FBT
and the high voltage generating circuit 40 including the flyback
transformer FBT according to the present embodiment are similar to
those shown earlier in FIG. 6, and therefore their description will
be omitted here.
[0091] The high voltage generating circuit 40 of the present
embodiment supplies the control circuit 1 with a divided voltage
output obtained by dividing the high direct-current voltage EHV by
a voltage dividing resistance R1-R2. The control circuit 1 changes
a detection output level according to the level of the supplied
divided voltage of the high direct-current voltage EHV. The
detection output is inputted to the oscillating circuit 4 on the
primary side via the photocoupler 2. The oscillating circuit 4
changes the frequency of the oscillating signal according to the
inputted detection output level, as described above, and then
supplies the oscillating signal to the driving circuit 4.
[0092] Accordingly, the switching frequency of the switching device
Q1 which is driven by driving voltage outputted from the driving
circuit 4 is changed, and thus switching output transmitted to the
flyback transformer FBT is controlled. As a result, irrespective of
variations in a load connected to the high direct-current voltage
EHV, for example, primary-side alternating voltage V1 at the
primary winding N1 of the flyback transformer FBT is controlled to
a constant level. Accordingly, the high direct-current voltage EHV
obtained on the secondary side of the flyback transformer FBT is
also controlled to a constant level.
[0093] Thus, the high voltage stabilizing circuit of the present
embodiment stabilizes the high direct-current voltage EHV.
[0094] FIGS. 2A to 2D are waveform diagrams showing switching
operation of the switching device Q1 in the high voltage
stabilizing circuit of FIG. 1.
[0095] The switching operation shown in FIGS. 2A to 2D is obtained
when the primary winding N1 in the high voltage stabilizing circuit
of FIG. 1 has the number of turns, 75T and an inductance L1 of 450
.mu.H, and each of the step-up windings NHV1 to NHV5 has the number
of turns, 500T. The parallel resonant capacitor Cr is set at 5600
pF and the direct current blocking capacitor C11 is set at 1 .mu.F,
while the inductance L2 of the choke coil CH is set at 50 .mu.H.
Load conditions are set so as to provide a high voltage load
current IHV=1 mA (at a high voltage load power of 27 W) to 0 mA (at
no load power of 0 W) at a high direct-current voltage EHV=27
KV.
[0096] FIGS. 2A and 2B show a switching output current IQ1 flowing
in the drain of the switching device Q1 and a parallel resonance
voltage VQ1 across the parallel resonant capacitor Cr,
respectively, at a commercial alternating-current power VAC=90 V
and at a high voltage load current IHV=1 mA. FIGS. 2C and 2D show a
switching output current IQ1 and a parallel resonance voltage VQ1,
respectively, at a commercial alternating-current power VAC=120 V
and at a high voltage load current IHV=0 mA.
[0097] First, as shown in FIG. 2A, in operation at a commercial
alternating-current power VAC=90 V and at a high voltage load
current IHV=1 mA (at a load power of 27 W), the switching output
current IQ1 is maintained at a zero level for a period TOFF for
which the switching device Q1 is turned off, and increases so as to
form a saw tooth-shaped line for a period TON for which the
switching device Q1 is turned on.
[0098] As shown in FIG. 2B, as a result of resonant operation of
the primary-side parallel resonant circuit that occurs during the
off period of the switching device Q1, the parallel resonance
voltage VQ1 forms a sinusoidal pulse waveform during the period
TOFF.
[0099] When the switching frequency of the switching device Q1 is
variably controlled, the period TOFF during which the switching
device Q1 is turned off within one switching period of switching
operation of the voltage resonance type converter is kept constant,
while the period TON during which the switching device Q1 is turned
on is variably controlled. The period TOFF is expressed as:
TOFF=.pi.{square root}{square root over (L1.multidot.Cr)} [Equation
1]
[0100] L1: inductance of primary winding N
[0101] Cr: capacitance of parallel resonant capacitor Cr
[0102] The inductance L1 (=450 .mu.H) of the primary winding N1 and
the capacitance (=5600 pF) of the parallel resonant capacitor Cr
are set as described above, and therefore according to the above
equation 1, the period TOFF remains 5 .mu.s regardless of whether
the switching frequency is variably controlled according to load
variations. This is also understood from FIGS. 2A and 2B and FIGS.
2C and 2D, in which both of the periods TON are 5 .mu.s.
[0103] The level of the parallel resonance voltage VQ1 in the
parallel resonant circuit is expressed as:
VQ1=Ei{1+.pi./2(TON/TOFF)} [Equation 2]
[0104] In the present embodiment, at a commercial
alternating-current power VAC=90 V and at a high voltage load
current IHV=1 mA, the period TON during which the switching device
Q1 is turned on is set at 25.5 .mu.s, as shown in FIGS. 2A and 2B.
Thus, a resonance pulse of the parallel resonance voltage VQ1
becomes 900 Vp. The parallel resonance voltage VQ1 includes a
rectified and smoothed voltage Ei=100 V that is obtained at a high
voltage load power of 27 W. Thus, a switching output voltage V1, or
an alternating voltage occurring in the primary winding N1, is 800
Vp, a voltage level obtained after the direct current blocking
capacitor C11 cuts off the rectified and smoothed voltage Ei=100 V
from the parallel resonance voltage VQ1=900 V. The switching
frequency fs in this case is controlled so as to ultimately become
33 KHz.
[0105] At a commercial alternating-current power VAC=120 V and at a
high voltage load current IHV=0 mA (at no load), the switching
frequency of the switching device Q1 is controlled to a higher
level. Practically, the period TON during which the switching
device Q1 is turned on is set to be TON=15 .mu.s, as shown in FIGS.
2C and 2D. The switching frequency fs in this case is fs=50
KHz.
[0106] As shown in FIG. 2C, the switching output current IQ1 is
maintained at a zero level during the period TOFF, and forms a saw
tooth-shaped waveform during the period TON. As shown in FIG. 2D,
the parallel resonance voltage VQ1 forms a sinusoidal resonance
pulse waveform during the period TOFF. To make sure of the above
description, the period TOFF is constant at 5 .mu.s in this case,
regardless of change in the switching frequency.
[0107] Since the switching frequency is set as described above on
an assumption that a rectified and smoothed voltage Ei=170 V is
obtained, the resonance pulse of the parallel resonance voltage VQ1
is at a level of 970 Vp. Thus, the switching output voltage V1
across the primary winding N1 is 800 Vp, a voltage level obtained
after the direct current blocking capacitor C11 cuts off the
rectified and smoothed voltage Ei=170 V from the parallel resonance
voltage VQ1=970 V.
[0108] Thus, the high voltage stabilizing circuit of the present
embodiment shown in FIG. 1 variably controls the switching
frequency according to variation in the high direct-current voltage
EHV caused by load variation or variation in alternating input
voltage, whereby the switching output voltage V1 obtained at the
primary winding N1 of the flyback transformer FBT is controlled to
a constant level of 800 Vp, for example. Hence, as described above,
the high direct-current voltage EHV obtained on the secondary side
of the flyback transformer FBT is also controlled to a constant
level of 27 KV, for example.
[0109] A comparison of the high voltage stabilizing circuit shown
earlier in FIG. 6 as a conventional circuit with the circuit of the
present embodiment shown in FIG. 1 indicates the following.
[0110] The circuit shown in FIG. 6 allows the switching frequency
of the switching converter for power conversion on the primary side
to be in synch with the horizontal synchronizing signal frequency
fH. Therefore, in order to stabilize the high direct-current
voltage EHV, it is necessary to variably control an on period
within one switching cycle (PWM control) under a condition where
the switching frequency is fixed to the horizontal synchronizing
signal frequency. Thus, two switching converters, that is, the
step-down converter 20 and the voltage resonance type converter 30
need to be provided on the primary side for power conversion.
[0111] On the other hand, the switching frequency of the voltage
resonance type converter provided on the primary side of the high
voltage stabilizing circuit of FIG. 1 is independent of the
horizontal synchronizing signal frequency fH and hence not in
synchronism therewith. Thus, it is possible to stabilize the high
direct-current voltage EHV by variably controlling the switching
frequency as described above. This means that it is possible to
stabilize the high direct-current voltage EHV by providing only one
voltage resonance type converter on the primary side of the high
voltage stabilizing circuit of the present embodiment.
[0112] Thus, power conversion efficiency in the circuit shown in
FIG. 1 is greatly improved. In actuality, at a high voltage load
power PHV=27 W, for example, power conversion efficiency in the
circuit shown in FIG. 6 is 72%, while that in the circuit shown in
FIG. 1 is improved to as high as 86%.
[0113] In addition, the alternating-current input power of the
circuit in FIG. 1 is reduced. At a high voltage load power PHV=27
w, for example, the alternating-current input power of the circuit
in FIG. 6 is 37.5 W, while that of the circuit in FIG. 1 is reduced
to about 31.4 W.
[0114] Moreover, as described above, the circuit shown in FIG. 1 is
provided with only one voltage resonance type converter on the
primary side, and therefore the number of parts for forming the
circuit is reduced, thereby making it possible to reduce size of
the circuit board and cost of the circuit.
[0115] Furthermore, the circuit shown in FIG. 1 makes it possible
to readily reduce the distortion of a white peak image described
with reference to FIG. 9 by effecting power supply at a maximum
load power in a state in which the switching frequency is decreased
to the lower limit of its control range, for example, in displaying
a white peak image. Thus, it is not necessary to increase
capacitance of the capacitors in the circuit, which also makes it
possible to reduce the size and cost of the circuit.
[0116] Another example of a high voltage stabilizing circuit
according to another embodiment of the present invention will next
be described with reference to a circuit diagram of FIG. 3. In FIG.
3, the same parts as in FIG. 1 are identified by the same reference
numerals, and their description will be omitted.
[0117] The high voltage stabilizing circuit shown in FIG. 3 is
provided with a self-excited voltage resonance type converter on
the primary side of a flyback transformer FBT. The present
embodiment also includes one switching device Q1, which performs
switching operation by the single-ended system. In this case, a
high voltage bipolar transistor (BJT; junction transistor) is used
as the switching device Q1.
[0118] Also in this case, a rectifying and smoothing circuit formed
by a bridge rectifier circuit Di and a smoothing capacitor Ci is
provided. The rectifying and smoothing circuit generates rectified
and smoothed voltage whose level substantially corresponds to an
alternating-current input voltage VAC multiplied by unity, and
supplies the voltage resonance type converter with the rectified
and smoothed voltage as direct-current input voltage.
[0119] A base of the switching device Q1 is connected to a positive
electrode side of the smoothing capacitor Ci (rectified and
smoothed voltage Ei) via a starting resistance RS, so that base
current at the start of power supply is taken from a line of the
rectifying and smoothing circuit. Connected between the base of the
switching device Q1 and a primary-side ground is a series resonant
circuit for self-oscillation driving that is formed by connecting a
driving winding NB, a resonant capacitor CB, and a base current
limiting resistance RB in series with each other.
[0120] A clamp diode DD inserted between the base of the switching
device Q1 and a negative electrode of the smoothing capacitor Ci
(primary-side ground) forms a path of clamp current that flows
during the off period of the switching device Q1. A collector of
the switching device Q1 is connected to one end of a primary
winding N1 of the flyback transformer FBT, while an emitter of the
switching device Q1 is grounded. Thus, switching output of the
switching device Q1 is transmitted to the primary winding N1 of the
flyback transformer FBT.
[0121] In this case, a parallel resonant capacitor Cr is connected
in parallel with the collector and emitter of the switching device
Q1. Capacitance of the parallel resonant capacitor Cr and leakage
inductance L1 of the primary winding N1 side of the flyback
transformer FBT form a primary-side parallel resonant circuit of
the voltage resonance type converter. Also in the self-excited
voltage resonance type converter, voltage V1 across the resonant
capacitor Cr forms a resonance pulse during the off period of the
switching device Q1, whereby voltage resonance type operation is
obtained.
[0122] An orthogonal type control transformer PRT is a saturable
reactor provided with a resonance current detecting winding ND, the
driving winding NB, and a control winding NC. The orthogonal type
control transformer PRT is provided to drive the switching device
Q1 and effect control for constant voltage.
[0123] The structure of the orthogonal type control transformer
PRT, not shown in the figure, is a cubic core formed by connecting
two double U-shaped cores each having four magnetic legs with each
other at ends of the magnetic legs. The resonance current detecting
winding ND and the driving winding NB are wound around two given
magnetic legs of the cubic core in the same winding direction, and
the control winding NC is wound in a direction orthogonal to the
resonance current detecting winding ND and the driving winding
NB.
[0124] In this case, the resonance current detecting winding ND of
the orthogonal type control transformer PRT is inserted in series
between the positive electrode of the smoothing capacitor Ci and
the primary winding N1 of the flyback transformer FBT, so that the
switching output of the switching device Q1 is transmitted to the
resonance current detecting winding ND via the primary winding N1.
The switching output obtained by the resonance current detecting
winding ND of the orthogonal type control transformer PRT is
induced in the driving winding NB via transformer coupling, whereby
an alternating voltage is generated as driving voltage in the
driving winding NB. The driving voltage is outputted as driving
current to the base of the switching device Q1 from the series
resonant circuit (NB and CB), which forms a self-oscillation
driving circuit, via the base current limiting resistance RB. Thus,
the switching device Q1 performs switching operation at a switching
frequency determined by the resonance frequency of the series
resonant circuit (NB and CB).
[0125] The structure and operation of the flyback transformer FBT
and a high voltage generating circuit 40 shown in FIG. 3 are the
same as described with reference to FIG. 1, and therefore their
description will be omitted here. However, also in this case,
alternating voltage generated in the primary winding N1 of the
flyback transformer FBT by the transmitted switching output of the
switching device Q1 is used to ultimately generate and output high
direct-current voltage EHV.
[0126] Also in the high voltage stabilizing circuit shown in FIG.
3, a voltage level obtained by dividing the high direct-current
voltage EHV by a voltage dividing resistance R1-R2 is inputted as a
detection voltage to a control circuit 1, whereby the control
circuit 1 detects variation in the level of the high direct-current
voltage EHV.
[0127] In this case, a tertiary winding N3 is provided on the
primary side of the flyback transformer FBT, and a half-wave
rectifier circuit comprising a diode D3 and a capacitor C3 is
connected to the tertiary winding N3, so that low direct-current
voltage is generated. The half-wave rectifier circuit is configured
so as to supply the control circuit 1 with the low direct-current
voltage as operating power.
[0128] The control circuit 1 in this case operates to change the
level of control current (direct current) flowing through the
control winding NC according to change in the high direct-current
voltage EHV. Then, inductance LB of the driving winding NB provided
in the orthogonal type control transformer PRT is variably
controlled by the control current. This changes resonance
conditions of the series resonant circuit including the inductance
LB of the driving winding NB within the circuit for
self-oscillation driving of the switching device Q1. Accordingly,
the operation of the control circuit 1 changes the switching
frequency of the switching device Q1.
[0129] As in the case of the circuit shown in FIG. 1, components in
the circuit are selected so that the switching frequency fs is
variably controlled in a required control range (for example fs=33
KHz to 50 KHz) according to variation in the high direct-current
voltage EHV and thereby switching output voltage V1 obtained at the
primary winding N1 of the flyback transformer FBT becomes constant.
Thus, the high direct-current voltage EHV is stabilized. A circuit
configuration such as to achieve the same operation as described
with reference to the waveform diagram of FIGS. 2A to 2D, for
example, suffices for actual switching frequency control in this
case.
[0130] The circuit configuration of FIG. 3 described so far can be
considered to be the circuit configuration of FIG. 1 in which the
choke coil CH and the direct current blocking capacitor C11 are
removed and the voltage resonance type converter is externally
excited. Such a configuration of the high voltage stabilizing
circuit as shown in FIG. 3 can be applied in cases where load
conditions of the high direct-current voltage EHV do not include
high load peak power, for example. As described above, the high
voltage stabilizing circuit may be configured with a simple
self-oscillation circuit and without the choke coil CH, the direct
current blocking capacitor C11 or the like, for example, and
therefore it is possible to further reduce size and cost of the
high voltage stabilizing circuit.
[0131] In the present embodiment, the orthogonal type control
transformer is used to effect control for constant voltage in a
circuit configuration provided with the self-excited resonance
converter on the primary side; however, an oblique type control
transformer proposed by the present applicant prior to the present
invention may be employed instead of the orthogonal type control
transformer.
[0132] As in the case of the orthogonal type control transformer,
the structure of the oblique type control transformer, which is not
shown in the figure, is a cubic core formed by combining two double
U-shaped cores each having four magnetic legs with each other, for
example. A control winding NC and a driving winding NB are wound
around the cubic core in obliquely crossing winding relation to
each other. Specifically, either the control winding NC or the
driving winding NB is wound around two adjacent magnetic legs of
the four magnetic legs of the cubic core, and the other winding is
wound around the two other magnetic legs situated in diagonal
positional relation to the above two magnetic legs.
[0133] The oblique type control transformer operates in such a
manner that inductance of the driving winding is increased even
when alternating current flowing through the driving winding is
changed from a negative current level to a positive current level.
Thus, the level of current in a negative direction for turning off
the switching device is increased, and thereby storage time of the
switching device is shortened. As a result of this, fall time of
the switching device at turn-off is also shortened, thereby
enabling further reduction of power loss in the switching
device.
[0134] In the circuit of the present embodiment, a self-excited or
externally excited voltage resonance type converter employing a
single-ended system is described as an example of a switching
converter to be provided on the primary side; for example, however,
a self-excited or externally excited voltage resonance type
converter employing a push-pull system that is provided with two
switching devices may be used.
[0135] The high voltage stabilizing circuit of the present
embodiment has been described on an assumption that it is applied
to a display unit for a personal computer; however, the high
voltage stabilizing circuit of the present embodiment can be
applied to, for example, television receivers and various other
display units having CRTs as display devices.
[0136] As described above, the high voltage stabilizing circuit
according to the present invention has a voltage resonance type
converter as a switching converter that performs switching
operation using direct-current input voltage obtained from
commercial alternating-current power, and switching output of the
voltage resonance type converter is directly transmitted to a
primary winding of a flyback transformer FBT (high voltage output
transformer). Then, a rectifier circuit for high voltage connected
on the secondary side of the flyback transformer FBT provides high
direct-current voltage as an anode voltage for a CRT, for
example.
[0137] The high direct-current voltage is stabilized by controlling
the switching frequency of the voltage resonance type converter on
the primary side according to the level of the high direct-current
voltage. In this case, the voltage resonance type converter on the
primary side has a free running switching frequency independent of
a horizontal synchronizing signal frequency, for example.
[0138] Such a configuration requires only one switching converter
to be provided on the primary side of the high voltage stabilizing
circuit. Thus, overall power conversion efficiency in the high
voltage stabilizing circuit is greatly improved, and also
alternating-current input power is reduced.
[0139] In addition, since the high voltage stabilizing circuit is
provided with only one switching converter on the primary side, the
number of parts for forming the circuit is greatly reduced, thereby
making it possible to reduce size of the circuit board and cost of
the circuit.
[0140] Moreover, the circuit configuration according to the present
invention makes it possible to readily and effectively reduce
distortion of a white peak image by lowering the switching
frequency to a required level and by increasing power supplied to a
load to a required level. Thus, it is not necessary to increase
capacitance of the capacitors in the circuit, which also makes it
possible to reduce the size and cost of the circuit.
[0141] While a preferred embodiment of the present invention has
been described using specific terms, such description is for
illustrative purposes only, and it is to be understood that changes
and variations may be made without departing from the spirit or
scope of the following claims.
* * * * *